WO2008003480A2 - Signal encoder, signal decoder, method for configuring a signal encoder, method for configuring a signal decoder, method for operating a signal encoder, method for operating a signal decoder, computer programme and data transfer system - Google Patents

Abstract

The invention relates to a signal encoder (110) for generating N' parallel binary signals (C0, C1, C2, C3, C4) based on B information bits (b0, b1, b2,b3). Said encoder comprises a vector mapping device for the non-linear mapping of possible bit patterns of the B information bits to 2B signal combinations on the N' parallel binary signals, where N' is greater than B. The signal encoder for generating a plurality of individual modulation symbols based on B information bits comprises a vector mapping device for the non-linear mapping of possible bit patterns of the B information bits to various 2B combinations consisting of at least two individual modulation symbols, the various 2B combinations representing a real subset of a complete modulation alphabet. Said complete modulation alphabet is formed by a cross product of sets of actual occurring modulation values of the two individual modulation symbols. The concept according to the invention permits the digital transfer of information using a redundant space code, allowing information to be reliably transferred even if a rough quantization exists.

Description

 Signal coder, signal coder, method of configuring a signal coder, method of configuring a signal coder, method of operating a signal coder

Signal encoder, method for operating a signal decoder, computer program and

Data transfer system

The present invention relates generally to a signal encoder, a signal decoder, methods of configuring a signal encoder, methods of configuring a signal decoder, methods of operating a signal encoder, methods of operating a signal decoder, a computer program and a data transmission system, in particular a redundant space Coding for digital communication over multi-channel systems with coarse quantization of the received signal.

Both in the context of wired and wireless data transmission, a digital communication is typically used today. The following briefly outlines the technical environment in which the present invention can be used. After a brief overview of a digital communication system in general, the module of digital modulation and the generation of the signal space constellation will be discussed, since, for example, the invention can be used in this area.

Fig. 12 shows a functional diagram and a block diagram of a digital communication system, respectively. The digital communication system of FIG. 12 is designated 1200 in its entirety. The digital communication system 1200 consists of the following nine essential components (see [I]):

1. Information source 1210: the information source 1210 provides the information to be transmitted in analog form (eg, an audio signal or a video signal) or in digital form (eg, a computer file).

2. Source coding 1220: the signals of the information source 1210 are converted by the source encoding 1220 in a stream of bits (also referred to as bitstream). As a rule, an attempt is made to remove any existing redundancy so as to represent the input signal with as few bits as possible (data compression).

3. Channel Coding 1230: Since the bitstream after source coding 1220 is susceptible to errors, error protection is often performed using channel coding. In this case, additional bits (so-called check bits) are added to the bit stream, which can be used on the receiver side to correct transmission errors.

4. Digital Modulation 1240: The channel coded bitstream (from channel encoding 1230) is mapped to waveforms that can be transmitted over the (always in practice) analog channel. In this case, a certain number B of bits is combined from the bit stream and then mapped to 2 ^B different signal forms.

5. Channel 1250: The channel 1250 transmits the waveforms and returns them usually in a distorted and noisy form at its output again. Technical transmission channels can be both wired (eg copper or fiber optic cable) and wireless (radio link) available. If the transmitter and receiver have several antennas, or if several cables of a cable are used simultaneously, a multichannel system is created (compare [2]). 6. Digital Demodulation 1260: The noisy, distorted waveforms (over channel 1250) are now mapped to a stream of numbers representing estimates of the transmitted encoded bitstream. The estimates usually include information about the reliability of the estimate. This is helpful in the subsequent channel decoding 1270.

7. Channel Decoding 1270: The estimates of the transmitted bit stream (obtained from digital demodulation 1260) are processed by a channel decoder, using the previously introduced check bits (compare channel decoding 1230) to correct for transmission errors. At the output of the channel decoder or the channel decoding 1270, a bit stream thus again arises which, in the case of successful error correction, exactly corresponds to the bit stream after the source coding 1220.

8. Source decoding 1280: The bit stream (obtained from the channel decoding) is decompressed again by a source decoder and, if necessary, returned to its original analog form.

9. Information sink 1290: The information sink 1290 is the receiver of the information, and receives the transmitted data from source decode 1280, for example.

In summary, it can thus be stated that the essential components of a digital communication system have been described with reference to FIG.

The digital modulation 1240 shown in FIG. 12 and the digital demodulation 1260 likewise shown with reference to FIG. 12 are explained in more detail below. In other words, the blocks "digital modulation" and "digital For this purpose, Fig. 13 shows a schematic representation of the digital modulation and the digital demodulation as well as a definition of an effective channel. In other words, Fig. 13 shows how a bit stream is mapped to waveforms.

The graph of Fig. 13 is designated in its entirety by 1300. The transmission system 1300 according to FIG. 13 receives a bit stream 1310, for example from the channel coding 1230 according to FIG. 12.

1. Signal space constellation 1320: In the context of the formation of a signal space constellation (also referred to as signal space illustration 1320), a group of B bits is mapped to one of 2 ^B possible complex (or real) numbers, the so-called signal space constellation. Detailed explanations with regard to the signal space constellation can be found in the further description.

2. Transmit signal processing 1330: If necessary, a number stream 1324 of real or complex numbers generated on the basis of the image on the signal space constellation is converted into a second number stream 1334 by so-called transmit signal processing. This is usually done to compensate for the channel (full or partial) distortions.

3. Pulse shaping 1340: The resulting (from the transmission signal processing) number stream 1334 is then converted by a pulse shaping 1340 into an analog transmission signal 1344. The pulse shaping 1340 serves to adapt the signal to the available bandwidth of the channel. 4. RF modulation 1350: If the channel (for example channel 1250, 1360) is a radio link, the signal must still be brought into the intended radio frequency range. This is done by mixing (eg, amplitude modulation) with a radio frequency (RF) carrier signal generated by a local oscillator. RF modulation 1350 also includes filtering and amplification of the RF signal. In the case of a wired channel, the RF modulation can also be omitted (so-called baseband transmission).

An RF signal 1354 generated by RF modulation 1350 is then transmitted over a wired or wireless channel 1250, 1360 (which is typically a multi-channel). A multiple channel allows the same transmission of multiple signals or signal components, which can lead to a coupling between the individual signals or signal components.

In the transmission over the channel 1360, a received signal 1364, which is supplied to a digital demodulation. In digital demodulation, estimates of the transmitted (channel coded) bits are generated from the received waveforms (the receive signal 1364).

The digital demodulation typically takes place in several stages, which are described below.

1. RF Demodulation 1370: If an RF modulation 1350 has been carried out on the transmitter side, the received signal typically only has to be mixed down again from the RF range into the baseband. In the case of baseband transmission, RF demodulation 1370 is omitted. 2. Analog-to-Digital Conversion 1380: Since modern signal processing is practically carried out only digitally, the analog reception values 1374 (typically supplied by the RF demodulation 1370) must be converted into a form that is both time- and value-discrete. This is done by means of an analog-to-digital converter 1380 (also referred to as A / D converter), or by means of a bank of A / D converters in a multi-channel system. The output of the analog-to-digital converter or bank of analog-to-digital converters can be interpreted as a stream 1384 of real or complex numbers.

3. Detection 1390: The mentioned 1384 may also be similar to the transmission signal processing at

Receiver be further processed. The goal is to produce estimates 1394 for the transmitted bit stream 1310.

It is further noted that the functionality of the transmit signal processing 1330, the pulse shaping 1340, the transmission over the channel 1360 as well as optionally further present RF modulation 1350 and RF demodulation 1370 will be understood in the following as transmission over an effective channel. In other words, the effective channel includes the transmit signal processing 1330, the pulse shaping 1340 and the transmission over the channel 1360. Further, the effective channel optionally also includes the RF modulation 1350 and the RF demodulation 1370. The effective channel thus describes the Transmission between the transmitter-side signal-space map 1320 and the receiver-side analog-to-digital conversion 1380 (which may also be described by a threshold decision).

14 therefore shows a schematic representation of a digital modulation and demodulation with an effective MIMO channel. Details regarding the effective MIMO channel are described below. The blocks "Transmit Signal Processing" 1330, "Pulse Forming" 1340, "RF Modulation" 1350, "Channel" 1360 and "RF Demodulation" 1370 may be combined into a single block with respect to the present invention This results in a block diagram shown in Fig. 14, which shows the mapping of the transmit-side bit stream 1310 to the receive-side stream 1394 of associated estimates, in other words, Fig. 14 shows an image of the transmit-side bitstream on the receive-side stream of its estimates.

It should be noted here that the same devices and signals in Figs. 13 and 14 are designated by the same reference numerals.

The effective channel 1360 has a certain number N of channel inputs at which it accepts complex or real numbers 1324 from the signal space constellation, and a number M of channel outputs at which analog signals 1374 can be tapped from the analog to digital converter 1380. Incidentally, such a channel with N inputs and M outputs is also called MN system or MIMO system (for multiple-input-multiple-output) (compare [2]). For example, a channel with N inputs and M outputs may result from the transmitter being equipped with N antennas and the receiver with M antennas. However, due to the transmit signal processing 1330 contained in the effective channel 1360, the number of transmit antennas may also be greater than N.

The present invention has no effect on the properties of the MIMO effective channel 1360. However, it is preferred that the characteristics of the MIMO effective channel 1360 be known. Hereinafter, Xi, X2, ..., X _N denote the numbers at the inputs of the effective channel 1360. Further, yi, Y _2r - _' - _f Y _M denote the numbers at the outputs of the effective channel 1360. It is assumed hereafter in that the properties of the effective channel 1360 are completely described by a so-called channel matrix H and a so-called noise vector n, where:

The receive signals yi, V ₂ ,..., YM 1374 from the outputs of the effective channel can therefore be represented as a linear combination of the transmit signals Xi, X2,..., X _N plus a noise term n. This is the case, inter alia, when the transmission channel 1360 is non-dispersive (frequency flat) in time, and the steps of signal processing 1330, pulse shaping 1340, RF modulation 1350, and RF demodulation 1370 are linear.

In the following, details are described with regard to the signal space constellation or signal space mapping. 15 shows a graphical representation of signal space mapping, MIMO effective channel transmission, and analog-to-digital conversion. The graphical representation of FIG. 15 is designated 1500 in its entirety. In other words, the mapping of the bit stream 1310 from the channel coding to the signal space constellation or signals 1324 is explained in more detail below. As sketched in FIG. 15, B bits Ci, C2,..., C _{B of} the bit stream are first of all used with the aid of of a serial / parallel converter (S / P) 1510 and then distributed to N signal space blocks sigi, sig ₂ , ..., sig _N. In this case, the i-th signal space block of the N signal space blocks 1520_1, 1520_2,..., 1520_N receives a number bi of the total of B bits and maps these to one of 2 ^bl possible complex (or real) numbers. The following mapping rule applies:

{- 1; l} ^{bi 2} ^> X ₁ e M ₁ .

In other words, B bits of the bit stream 1310 of the channel coding are divided into N groups of bits, with an i th group comprising bi bits. The i-th signal space block sigi (also denoted 1520_i) thus receives the group of bi bits and maps them to a real or complex value Xi (where preferably: 1 <i <N).

The number Xi is chosen from a 2 ^1- element set M ±, where the set Mi is called a modulation alphabet (or a single modulation alphabet) (for a group of bi bits). Incidentally, the number Xi is also referred to as a single modulation symbol. If the numbers Xi, X ₂ , • •. , X _{N to} an N-dimensional vector x according to

x = [X ₁ X ₂ ... x _N ] ^τ e M

summarized, the N-dimensional vector x is an element of the set M, which contains all possible combinations of the elements of the sets Mi, Mz, ..., M _N. In other words, it holds:

M = M ₁ XMΣ X. • • XM _N.

Further away

bi + b ₂ + ... + b _N = B holds, the set M (also referred to as the total modulation alphabet) has exactly \ M \ = 2 ^B elements. In this way, by the group of B bits from the bit stream 1310, exactly one vector x out of the 2 ^B possible vectors from M is selected.

The following describes the analog-to-digital conversion following transmission over the effective MIMO channel. The effective MIMO channel 1360 receives it as inputs the individual modulation symbols xχ _f x _2, ..., x _N and the vector x. The effective MIMO channel 1360 generates received signals yi, V2, ..., yw from the vector x M. Further, noise occurs in transmission over the effective MIMO channel 1360. Thus, after transmission via the effective MIMO channel 1360, a noisy receive vector y is obtained at the output of the MIMO channel 1360, where:

y = Hx + n.

The noisy receive vector y is then fed to a bank 1540 of A / D converters 1540_1, 1540_2, ..., 1540_M. In other words, the ith analog-to-digital converter 1540_i of the analog-to-digital converter bank 1540 receives the i-th receive signal yi. At an output of the i-th A / D converter 1540_i is thus an ri bit wide number z ± available, which corresponds to the quantized signal or received signal y ±:

Zi = Qi (yi),

The function Qi (.) Describes the operation of the ith A / D converter. The total resolution of all A / D converters is given by r _ges = ri + r ₂ + ... + r _M bits. So that it is even possible to reconstruct the B bits sent, the following must apply:

r _ges ≥ B. In the following, the technical problem underlying the present invention will be described. Thus, the interconnections between the channels of a multi-channel system (eg, an MIMO effective channel 1360) make it difficult to realize high-resolution A / D conversion. In practice this can lead to the fact that with technically available A / D converters, no A / D conversion can be made, which would be high-resolution enough to implement relevant signal processing methods (compare [3]) and channel coding (cf. [1]).

In the following, the implementation of an analog / digital conversion (A / D conversion) is described in multi-channel systems. Relevant methods of transmit and receive signal processing and channel coding refrain from the existence of A / D converters at a receiver, assuming that the continuous signal values at the output of the effective channel are available for further signal processing and channel decoding stand. This assumption is justified if the resolution of the A / D converters used is so high that the generated quantization noise is negligible. The coupling between the channels of a multi-channel system, however, ensures that it is technically difficult to realize such a high resolution of the A / D converter. The following is an example of a multichannel system described by the following channel matrix H of an effective channel:

- 31 - 81 23 - 42 75 21 - 49 - 102

1 - 118 84 12 154 4 - 94 61 40

H =

100 9 84 - 13 - 10 100 - 7 51 89 - 28

20 51 8 - 59 115 - 113 49

The system considered corresponds to the system explained with reference to FIG. 14. The effective channel 1360 thus has N = 4 channel inputs and Af = 4 channel outputs. Let the signal space constellation be a 4Q & M popular in practice, where for the modulation alphabets Mi, M ₂ ,, M ₃ , Mi:

Details regarding the signal space constellation have been explained above. The signal space constellation described leads to a modulation alphabet with 4 ⁴ = 256 elements, from which one is selected for transmission by a group of B = 8 bits of the bit stream 1310 (cf. FIG. 15 and the associated description).

In a noise-free case, one of 256 possible signals or one of 256 possible signal values (in the sense of a signal combination) then appears at the outputs yi, y ₂ , V ₃ , y _{4 of} the effective channel 1360.

Fig. 16 shows all 256 possible signals at the four outputs yi, y ₂ , Y3, Yi of the effective channel 1360. In other words, Fig. 16 shows a noiseless signal constellation at the outputs of the effective channel 1360 according to the aforementioned example. Horizontal and vertical lines indicate the limits of quantization intervals when using 2-bit A / D converters per dimension or per signal component.

In other words, Fig. 16 shows the output values appearing at the first output (eg at the output yi), at the second output (eg output y ₂ ), at the third output (e.g. At the output y ₃ ) and at the fourth output (eg at the output y _t ) when 256 different input vectors are applied to the inputs Xi, X ₂ , X ₃ , X4 of the effective MIMO channel 1360 or input modulation symbols (consisting of four individual modulation symbols from the sets Mi, M ₂ , M ₃ , M ₄ ) are created. It is here It should be noted that according to the channel matrix H described above, a coupling between all input signals X ₂ , X ₂ , X ₃ , X _{4 of} the effective MIMO channel 1360 and all outputs yi, y ₂ , y ₃ , Yi of the effective MIMO channel 1360 occurs. For this reason, considerably more than four different signal values are calculated for all output signals yi, y _2, Yz, Y _A (typically even up to 256 different signal values) while without Verkopp- lung between the individual inputs or channels only four different signal values would result.

For the function of relevant methods of signal processing and channel coding, it is necessary that all received signals (in the noise-free case) can be resolved by the A / D converters, i.e. that different signals are also assigned to different quantization intervals of the A / D converters. However, the dense proximity of various receive signals to be recognized in FIG. 16 forces (for this purpose) the use of relatively high resolution A / D converters. By increasing the number of channel inputs, or by increasing the number of stages of the modulation alphabet, the required resolution of the A / D converters quickly increases beyond the technical limits. As a result, high-resolution A / D conversion can not be performed for multi-channel systems above a certain size. As a result, so-called coarse quantization occurs.

The problem of coarse quantization is explained in more detail below. For this purpose, Fig. 16 is again used. Fig. 16 shows, for example, in a first graphical representation 1610 values of a received signal yi which is applied to an input of an A / D converter, for 256 different combinations of signal values of the transmission signals xl, _{X2, X3,} X4 according to the modulation alphabet AT. The different signal values of the received signal yi are, for example, as points in a complex number plane, that is to say divided according to real part and imaginary part, plotted. Here, for example, values in a range between -3 and 3 are taken into account for a real part, and values in a range between -3 and 3 for an imaginary part.

Further, the first graphic image 1610 shows as vertical lines interval boundaries of quantization intervals of an analog-to-digital converter, which has a real part of the signal yi in four stages (between -3 and -1.5, between -1.5 and 0, between 0 and 1.5 and between 1.5 and 3). Horizontal lines also show interval limits of quantization intervals of another analog-to-digital converter, which has an imaginary part of the received signal yi in four stages (between -3 and -1.5, between -1.5 and 0, between 0 and 1.5 and between 1.5 and 3).

A corresponding second graphic representation 1620 describes signal values of a received signal y _Σ as well as interval limits of two associated analog / digital converters (for a real part and for an imaginary part). Furthermore, a third graphical representation 1630 describes signal values of the third received signal y ₃ as well as interval limits of two associated analog / digital converters for quantizing the real part and the imaginary part of the third received signal y ₃ . A fourth graph 1640 further shows signal values of the fourth received signal y * including interval boundaries of two associated analog-to-digital converters for quantizing the real part and the imaginary part of the fourth received signal y _t .

In the graphs 1610, 1620, 1620 and 1640 according to FIG. 16, quantization interval limits which result in the case of a homogeneous quantization with 2-bit A / D converters are entered by way of example. In other words, a 2-bit A / D converter for the real part and a 2-bit A / D converter for the imaginary part respectively become each of the four outputs yi, y ₂ , y ₃ , Yi of the effective MIMO channel 1360 is connected

Each output of the channel 1360 is a coarse A / D conversion because different input signals are mapped to one and the same quantization interval. It turns out that even then a coarse quantization is involved if all four quantized outputs (zi, Z ₂ , Z ₃ , Z ₄ ) are considered together. In order to recognize this, two receive signals (ie two signals belonging to two different transmit signals) are highlighted in FIG. 16 by large "* ^w " markers 1650. It can be seen that at both four outputs both signals are mapped to the same quantization interval, and thus can not be distinguished. This loss of signal separability after quantization is the essential characteristic of coarse quantization.

In other words, in a coarse quantization, two different transmit signals belonging to two different information to be transmitted (or bit patterns of the information bits) result in equal signals at the outputs of receive-side A / D converters.

The following describes the effects of coarse quantization communication. The above-described uniqueness of various transmitted signals caused by coarse quantization ensures that relevant methods of signal processing and channel coding fail their service. To illustrate this, in one example, a convolutional code (memory length 6) with Viterbi decoding was adopted as the channel coding method, and so-called log-likelihood detection was adopted as received signal processing (see [4]). Fig. 17 is a graph showing a bit error rate after Viterbi decoding of a convolutional code. The graph of FIG. 17 is designated 1700 in its entirety. At an abscissa 1710, a ratio between a transmission power and a noise power in decibels is plotted. An ordinate 1720 shows a coded bit error rate in a logarithmic representation. A first curve 1730 writes a relationship between the signal-to-noise ratio (Pr / o ^) and the coded bit error rate using a 1-bit A / D converter in the evaluation of the received signals y _lf y ₂ , y ₃ , Y4. A second curve 1740 describes the same relationship for the use of an ∞-bit A / D converter, ie an A / D converter with arbitrarily high resolution. In other words, Fig. 17 shows the bit error frequencies resulting using the example channel as a function of the ratio of applied transmit power P _τ to power σ "of the receive noise. With a hypothetical, infinitely high resolution of the A / D converters, the coding system works excellently. However, when A / D converters with 1-bit resolution per dimension (ie, 1-bit resolution for the real part and 1-bit resolution for the imaginary part of the signals yi, y2, y3, yt) are used, the bit error rate saturates to one considerably high value of about 10 ^{~ 4} . This is too high for many practical applications. Therefore, the previously described encoding system can not readily cooperate with coarse A / D conversion (eg, with a 1-bit A / D converter).

Fig. 18 also shows a schematic representation of a bus system with four parallel lines. In the bus system shown, four bits (b ₀ , bi, b ₂ , b ₃ ) e {0, l} ^{4 are} to be transmitted. For this purpose, a so-called bus driver 1810 (consisting of four individual line drivers 1810_0, 1810_l, 1810_2, 1810_3) converts the binary values of the bits bo, bi, b _Σf b ₃ into one of two possible voltage values. For example, the first line driver 1810_0 the bit bo in an associated transmission voltage U ₀ . Line driver 1810_l converts bit bi to an associated transmit voltage ui. The line driver 1810_2 converts the bit b ₂ into an associated transmission voltage U ₂ . The line driver 1810_3 converts the bit b ₃ into an associated transmission voltage U ₃ . In the following, it is assumed that the transmission voltages _O i are: Q

1

The modulation alphabet of the bus system shown is therefore:

M = {0V, 5V} ⁴ .

The modulation alphabet of the bus system shown therefore consists of \ M \ = 16 different 4-dimensional vectors. The coupling of the lines (of the bus system) is described by the following, diagonally dominant, effective MIMO channel matrix:

1 0.35 0.1225 0.0429

0.35 1 0.35 0.1225

H =

0.1225 0.35 1 0.35

0.0429 0.1225 0.35 1

In this case, the lines are those elements of the bus system which assume the transmission of the transmission voltages Ui from the output of the transmitter to the input of the receiver, so that the reception voltages yo, yi, Yz, Y ₃ arise. At a receiver or at the receiving end of the bus system, the four received voltages yo, yi, y ₂ , y _{3 are} again assigned values from the set {0, 1}. In other words, reception signals or reception voltages yo, Yi Y _f Y _{3 2r} a threshold value 1830 are conces- leads, for example, four single threshold decision 1830_0, 1830_l, 1830_2, 1830_3 includes.

For example, the threshold decision 1830_0 receives the receive signal yo and generates therefrom an estimated receive bit b ₀ . Similarly, the single threshold decision 1830_l receives the receive signal yi and generates therefrom an estimated receive bit

, The single threshold value 1830_2 receives the received signal y2 and generates an estimated reception bit _B2. The Einzelschwellwertentscheider 1830_3 receives the received signal y ₃ and generates therefrom an estimated receive bit

In other words, at the receiving end of the bus system, the four received voltages (or receive signals yo, yi, y ₂ , y ₃ ) are again assigned values from the set {0, 1} (by the threshold decision 1830). The following mapping rule or assignment rule is used:

fθ if Y ₁ <2.5V b, = <

[1 otherwise

If all of the 16 possible transmit vectors consisting of four transmit voltages Uo, U ₁ , U ₂ , U _{3 are} transmitted from the modulation alphabet M in succession, the detected bits (b ₀ , JD ₁ , b ₂ , b ₃ ) are determined. 19 shows a table in which estimated receive bits (S ₃ , b ₂ ,

B ₀ ) bits (b ₃ , b ₂ , bi, b ₀ ) are compared for the 16 different combinations of transmitted bits (S ₃ , b ₂ , S ₁ , S ₀ ). In other words, FIG. 19 shows a comparison of transmitted and received bits of the bus system according to FIG. 18.

If, in turn, all of the 16 possible send vectors are transferred from the modulation alphabet M, then one finds for the detected bits (b ₀ , B ₁ , B ₂ , b ₃ ) have the values included in the table of FIG. 19. As can be seen from the table of FIG. 19, sixteen errors occur in the 16 × 4 = 64 transmitted bits. (Which will transfer the bits bi and b ₂₎ at the inner leads contact each two errors, and wherein the outer leads (on which the bits bo and b are transferred ₃₎ an error occurs in each case. This is because the two outer lines have only one dominant interfering adjacent line and therefore suffer less crosstalk than the inner lines. In any case, according to the prior art, a reliable transmission is not possible because the crosstalk is too pronounced, so that bit errors occur.

In view of the known state of the art, it is the object of the present invention to provide a concept for encoded transmission of digital data which, even in the presence of crosstalk between several transmission channels, reliably reconstructs the transmitted digital information using threshold value deciders Form of analog / digital converters finite or in particular low resolution allows.

This object is achieved by a signal encoder according to claim 1 or claim 10, by a method of configuring a signal encoder according to claim 25, 32, 71 or 72, by a signal decoder according to claim 37 or claim 44, a method of configuring a signal decoder according to claim 52, 57 or 81, a method for operating a signal encoder according to claim 62 or claim 63, a method for operating a signal decoder according to claim 64 or claim 65, a computer program according to claim 66 or a data transmission system according to claim 67 or 69.

The present invention thus provides a signal encoder according to claim 1. It is the gist of the present invention that by using a vector imager to nonlinearly map possible bit patterns of B information bits to 2 ^B signal combinations on the N 'parallel binary signals, with N' greater B, in a signal encoder achieves redundant space coding can be. The redundant space coding enables a reliable receiver-side detection of the N 'binary signals after a transmission via a channel, even if the channel has a crosstalk.

In other words, out of the 2 ^{N '} possible signal combinations that can be transmitted through the N' parallel binary signals (or over N parallel lines), only a total of 2 ^B signal combinations are used. The nonlinear vector imager thus serves to map the bit patterns of the B information bits, which the vector imager receives, for example, over B parallel lines, to 2 ^B signal combinations, so that exactly one signal combination is assigned to each of the bit patterns of the B information bits. Thus, the signal encoder according to the invention makes it possible to transmit only those of the 2 ^{N '} possible signal combinations on the N' parallel lines, which are distinguishable at a receiver even in the presence of crosstalk.

The present invention offers the significant advantage over conventional solutions that even in the presence of crosstalk between the N 'parallel binary signals of signals received at a receiver and signal decoder, it can still be reliably deduced to the transmitted B information bits while doing so without the use the redundant space coding according to the invention would not be possible.

The present invention further provides a signal encoder for generating a plurality of single modulation symbols. len based on B information bits, according to claim 10.

The signal coder according to claim 10, as well as the signal coder according to claim 1, realize the core idea that it is advantageous to map bit patterns of B information bits to 2 ^B different combinations of a larger overall modulation alphabet and overall signal space, respectively. Thus, not all possible combinations of single modulation symbols that are part of a total modulation alphabet are used for the transmission of the B information bits. Rather, the use of a true subset of combinations of single modulation symbols from an overall modulation alphabet allows only those combinations of single modulation symbols to be used at the transmitter end, even after transmission over a channel where crosstalk occurs between the transmitted single modulation signals to distinguish for sure.

Thus, as a whole, redundant space coding is achieved by the inventive concept in that of the available combinations of the individual modulation symbols only a real subset is used, which leads to reliable reception results.

The present invention further provides a method of configuring a signal encoder according to claim 25. The method of configuring the signal encoder realizes the gist of the present invention to adjust the mapping scheme of the vector image in the signal encoder such that the vector image includes the 2 ^B possible bit patterns of the B information bits 2 ^B signal combinations that lead to reliably distinguishable received signals. Based on the above concept, configuring the signal encoder includes determining receive signals at a receiver for more than 2 ^B different signal combinations of the N 'parallel signals at an output of the signal encoder to have a transmission result, and setting the mapping rule based on the transmission result. By knowing the received signals for more than 2 ^B different signal combinations of the N 'parallel signals at one output of the signal coder, it can be determined which 2 ^B different signal combinations from the more than 2 ^B different signal combinations to the best (ie for example best distinguishable ) Carry received signals. According to the transfer result, the vector image can thus be optimally configured.

Incidentally, a similar concept is realized by the method of configuring a signal encoder according to claim 32. The method according to claim 32 substantially corresponds to the method according to claim 25, wherein combinations of single modulation symbols are used instead of signal combinations of (binary) signals.

The present invention further provides a signal decoder for generating B information bits from N 'parallel signals according to claim 37. The signal decoder according to claim 37 realizes the concept of using redundant spatial coding in a digital communication system. Since more signal combinations of the N 'parallel signals are possible than there are bit combinations for the receive bits, several of the signal combinations on the N' parallel signals are mapped to the same combination of the B receive bits. It can thereby be achieved that a transmission system with a signal decoder according to the invention is particularly insensitive to channel-related interference. Thus, the signal decoder according to the invention makes it possible to map such signal combinations to identical bit combinations of the B receive bits, which can be assumed (for example with a predetermined minimum probability) to be parallel to the N 'on the same transmitted signal combination. based on signals. In other words, both an undisturbed signal combination based on a particular combination of B transmitted bits of information and a disturbed signal combination based on the same B transmitted bits of information can be mapped to the correct combination of receive bits. The fact that the vector image maps a greater number of states (the signal combinations or threshold valued signal combinations) on the N 'parallel signals to a smaller number of states (the possible combinations of the B receive bits) provides particularly high flexibility in the Selection of the vector image and thus enables the achievement of increased interference immunity by suitable choice of the vector image.

The signal coder according to claim 44 is based on the same core idea as the signal coder according to claim 37, wherein the signal combinations on the N 'parallel signals are replaced by combinations of single modulation symbols. A signal space at an input of the vector image includes more possible combinations of single modulation symbols than there are combinations of the B information bits. Thus, there is a redundant space coding at the input of the signal coder in the sense that not all available combinations of individual modulation signals are exploited for an information transmission.

The present invention further provides a method of configuring a signal decoder according to claim 52. The method comprises transmitting at least 2 ^B different signal combinations to the signal decoder, analyzing the received signals provided at an input of the signal decoder, an analysis result and setting a mapping rule depending on the analysis result. The inventive method for configuring a signal decoder is based on the principle that it is advantageous to have a mapping scheme of a vector image for mapping signal combinations on N 'parallel binary signals to combinations of B information bits, with N' greater B, by analyzing the received signals at an input of the signal decoder to set at least 2 ^B different signal combinations. By evaluating the received signals for the 2 ^B different signal combinations, 2 ^B partial mapping specifications (which describe an image of a combination of received signals onto a bit pattern of received information bits) can be determined from a total of 2 ^{N '} partial mapping specifications. Thus, in the manner according to the invention, the essential parts of the mapping rule describing a mapping of valid signal combinations on the N 'parallel (ideally binary) signals to the combinations of B information bits can be specified. Incidentally, the other parts of the mapping rule, which for example define an image of invalid or disturbed signal combinations of the N 'parallel signals on combinations of the B information bits, can likewise be set on the basis of the received signals in order to ensure, for example, the lowest possible error probability.

The present invention further provides a method of configuring a signal decoder according to claim 57. The method according to claim 57 is substantially analogous to the method according to claim 52, wherein combinations of single modulation symbols in place of signal combinations on the N 'parallel ones Signals occur.

The present invention further provides a method of operating a signal decoder according to claims 62 and 63, and methods of operating the signal decoder of claims 64 and 65. The present invention provides Furthermore, a computer program according to claim 66. Furthermore, the present invention provides a data transmission system according to claim 67, which combines the advantages of the signal coder according to the invention with the advantages of the inventive signal decoder.

Other preferred embodiments of the present invention are otherwise defined by the dependent claims.

Preferred embodiments of the present invention will be explained in more detail below with reference to the accompanying drawings. Show it:

1 is a block diagram of a bus system according to a first embodiment of the present invention;

FIG. 2 is a block diagram of a bus system according to another embodiment of the present invention; FIG.

Fig. 3 is a schematic representation of transmission lines for use in a bus system according to Fig. 1 or 2;

Fig. 4a is a schematic representation of a memory for use as a vector image in a signal decoder;

Fig. 4b is a schematic representation of a memory for use as a vector image in a signal encoder;

5 is a tabular representation of an image of bit patterns to be transmitted on signal combinations on a plurality of parallel (ideally binary) signals, and an image applying received signal combinations on a plurality of parallel (ideally binary) signals to reconstructed information bits;

Fig. 6a is a schematic representation of an original transmission system using digital modulation and demodulation;

6b shows a schematic representation of a digital transmission system according to the invention using digital modulation and demodulation, according to an embodiment of the present invention;

Fig. 7a is a schematic representation of an original digital transmission system using digital modulation and demodulation;

7b shows a schematic representation of a digital transmission system according to the invention using digital modulation and demodulation, according to an exemplary embodiment of the present invention;

8 is a schematic diagram of a digital transmission system using digital modulation and demodulation, according to one embodiment of the present invention;

9 is a graph showing a bit error rate after Viterbi decoding of a convolutional code, with and without using the inventive concept;

10a shows a flow chart of a method according to the invention for setting a mapping rule of a vector image in a signal coder. based on a transmission result, according to an embodiment of the present invention;

10 b is a flow chart of a method according to the invention for setting a mapping rule of a vector image in a signal coder based on a transmission result, according to an exemplary embodiment of the present invention;

10c shows a flow chart of a method according to the invention for setting a mapping rule of a vector image into a signal coder, according to an embodiment of the present invention

Invention;

10d shows a flow chart of a method according to the invention for setting a mapping rule of a vector image in a signal coder, according to an exemplary embodiment of the present invention;

11 a shows a flowchart of a method according to the invention for setting a mapping rule of a vector image in a signal decoder, according to an exemplary embodiment of the present invention;

11b shows a flow chart of a method according to the invention for setting a mapping rule of a vector image in a signal decoder, according to an embodiment of the present invention;

Fig. 12 is a schematic illustration of essential components of a digital communication system; Fig. 13 is a schematic diagram of a transmission system using digital modulation and demodulation, and an effective channel definition;

14 is a schematic representation of a digital transmission system using digital modulation and demodulation with an effective MIMO channel;

FIG. 15 shows a schematic representation of a digital transmission system with a signal space map and an analog / digital conversion; FIG.

Fig. 16 is a graph of signal constellations at the outputs of an effective channel including quantization intervals;

Fig. 17 is a graph showing a bit error rate after Viterbi decoding of a convolutional code;

Fig. 18 is a schematic diagram of a conventional bus system; and

19 shows a tabular comparison of combinations of transmitted information bits and combinations of received information bits in an original bus system according to FIG. 18.

1 shows a schematic representation of a bus system according to the invention according to a first exemplary embodiment of the present invention. The bus system according to FIG. 1 is designated in its entirety by 100. The bus system 100 includes a signal coder 110, which is designed to four information bits bo, bi, _b2, b3 to receive, and based thereon to five binary signals Uo, Ui, U2, U3, _U4 to produce. The transmission system 100 further includes a signal decoder 120, which is designed to receive five reception signals yo _f yi, y _2f y _{3 /} y _4, based on the Gesen ^¬ Deten binary signals, and based on the five received signals four received or reconstructed or estimated information bits b ₀ , S ₁ , b ₂ , b ₃ . The transmission system 100 further comprises a channel 130, which connects outputs of the signal coder 110 to inputs of the signal decoder 10. The channel 130 comprises, for example, five parallel lines, wherein a first line 130_0 connects a first output of the signal coder, to which the first output signal Uo is applied, to a first input of the signal decoder 120, to which the received signal yo is applied. In general, an ith line 130_i connects an output of the signal encoder 110, to which the output signal Ui is applied, to the input of the signal decoder 120, to which the receive signal or input signal yi is applied. On the channel 130 there is a crosstalk between different of the lines 130_0, 130_l, 130_2, 130_3, 130_4, which is characterized by corresponding lines, and which is based, for example, in a capacitive or inductive coupling of the lines.

The signal encoder 110 comprises a vector imager 140 and optionally a line driver 142. The vector imager 140 receives the four information bits bo, b _x , b ₂ , b ₃ and generates thereon five binary signals Co, Ci, C2, C ₃ , C ₄ . The signal combinations of the five binary signals Co, Ci, C2 _/ C ₃ , C ₄ form a redundant room code, as will be explained below. The optional line driver 142 receives the data provided by the vector images 140 five binary signals Co, Ci, cΣ, _C3, C4 and generated based thereon uo the output signals Ui, U2, U3, _U4 of the signal encoder 110 (also referred to as transmission signals) , The line driver 142 is configured to form an output Ui of the signal encoder 110 based on exactly one output signal c ± of the vector imager 142. In In other words, the optional line driver merely represents a driver or amplifier consisting of a plurality of mutually independent individual drivers 142_0, 142_1, 142_2, 142_3, 142_4, which preferably act as single-channel drivers with exactly one input and exactly one output ,

In the event that the optional line driver 142 is omitted, moreover, the output signals C ₀ , C _\ , C ₂ , C ₃ , C ₄ directly form the output signals u ₀ , Ui, U ₂ , U ₃ , u _{4 of} the signal encoder 110, where:

Ci = Ui (0≤i≤4).

It should be noted, moreover, that the signals Co, Ci, C ₂ , C ₃ , C ₄ and u <j, uj., U ₂ , U ₃ , U ₄ are hereinafter referred to as binary signals, since these in one Ideally, each assume exactly two different logical states. In a practical realization, it is in the signals C i, C _2, C _3, C _4, uo, Ui, U _2, U _3, U _4, however, typically signals are represented by voltage values or current values, that is to say in principle analog signals. The voltage or current values of said signals, however, usually assume values that are sufficiently close to predefined ideal values, by which two different logic states are described. Therefore, these signals can be interpreted as binary signals.

The signal decoder 120 includes an optional threshold detector 150 and a vector mapper 152. The optional threshold detector 150 receives the five receive signals yo, yi, y ₂ , y ₃ , y4 and generates five binary or threshold valued signals z o, z _lr z ₂ based thereon , z ₃ , Z ₄ , which describe the levels received by the signal decoder 120 on the lines 130_0, 130_l, 130_2, 130_3, 130_4. Threshold decision 150 typically consists of five different from each other in terms of NEN information signals independent of each individual threshold value decision 150_0, 15O-I, 150_2, 150_3, 150_4. An i th threshold decision 150_i receives the received signal yi and generates based on the binary signal Z ₁ . A threshold decision 150_i is generally designed to set the signal zi to a logic value of "0 ^M when the voltage of the input signal yi is less than a predetermined threshold, and to increase the signal Zi to a logical value of" 1 ^Λ> when the voltage of the input signal yi is greater than the predetermined threshold value. Alternatively, the threshold decision 150_i may also operate in a reverse manner. The threshold decision 150_i may thus be designed, for example, to assign a logic value of "1" to the signal Zi when the voltage of the input signal yi is less than a predetermined threshold value, and to assign a logical value of "0" to the signal Zi when the voltage of the input signal yi is greater than the predetermined threshold value.

It is preferred that the thresholds be the same for all of the five threshold decision gates 150_0, 150_l, 150_2, 150_3, 150_4, for example. Alternatively, however, the individual threshold value discriminators can also have different threshold values. Incidentally, it should be noted that the individual threshold decision makers may comprise, for example, a CMOS gate, of which at least one input is supplied with the receive signal yi. The threshold value is determined, for example, by the threshold voltages of the transistors of the CMOS gate. The corresponding signal Zi is derived in this case from the output of the CMOS gate. Alternatively, the single threshold decision 150_i may also include differential amplifiers in which a first input receives the respective input signal yi, and in which a second input receives a reference signal. The associated signal Z ₁ can then be derived from the output of the differential amplifier. It should be noted here that in the context of the present invention, a threshold value decider generally includes both a binary threshold value discriminator (with, for example, a threshold value or a hysteresis) for distinguishing two states and a threshold value decider with at least two threshold values for distinguishing more than two states , So for example, an analog-to-digital converter is understood. Examples of possible threshold decision makers are therefore a 1-bit decision maker, a 2-bit decision maker and an N-bit decision maker. More generally, the term thresholder includes, for example, a quantizer of arbitrary resolution, for example, configured to convert a value-continuous signal into information about which of a plurality of discrete value intervals or ranges of values there is a value of the continuous-value signal.

The vector imager 152 receives the threshold valued signals z ₀ , Zi, Z ₂ , Z ₃ , Z ₄ and maps them to received or reconstructed information bits b ₀ , S ₁ , b ₂ , B ₃ , respectively. The vector image is designed to perform the mapping in such a way that the received or reconstructed information bits b ₀ , S ₁ , b ₂ , S _{3 are present} at least in the presence of a noise-free transmission between the signal encoder 110 and the signal decoder 120 (FIG. but in the presence of crosstalk between the lines 130_0, 130_l, 130_2, 130_3, 130_4) the transmitted information bits bo, bi, b2, b ₃ correctly reproduce.

It is further noted that in the case that the threshold value is omitted 150 Empfangssigna- Ie y _0, y i, V _2, Yi, Vi directly the input signals Z _0, z _x, z _2, Z _3, Z ₄ of the Vector images 152 form. Based on the above structural description, the operation of the transmission system 100 will be explained in detail below.

The vector imager 140 in the signal encoder 110 generally receives B information bits per time step (typically parallel) and generates N 'binary signals Co, Ci, C ₂ , C ₃ , C ₄ per unit time N' representing the N 'output signals Uo, Ui, U ₂ , U ₃ , U _{4 of} the signal encoder 110, or which are mapped by a line driver 142 according to a one-to-one mapping on the output signals uo, Ui, U2, U ₃ , U _{4 of} the signal encoder 110. In a time interval or time step, therefore, there is one (or exactly one) of 2 ^B different combinations of information bits bo, bi, b ₂ , b ₃ to be sent. In a time interval or time step is also a (or exactly one) out of a total of 2 ^{N '} different combinations of the output signals U ₀ , U ₁ , U ₂ , U3, U _{4 of} the signal encoder 110, wherein the signal combinations of the output signals u ₀ , Ui, U2, U ₃ , u _{4 of} the signal encoder to the N 'parallel lines 130_0, 130-1, 130-2, 130_3, 130_4 as input signals or transmission signals are supplied. It should also be noted that the following applies:

N '> B.

Thus, not all of the total of 2 ^N 'possible signal combinations of the output signals Uo, Ui, U2, U ₃ , U _{4 of} the signal encoder 110 is associated with a combination of the information bits bo, bi, b ₂ , b3.

In a preferred embodiment, each combination of the information bits bo, bi, b2, b3 is assigned exactly one signal combination of the output signals u ₀ , Ui, U ₂ , U ₃ , U _{4 of} the signal coder 110. In other words, the vector image 140 forms the 2 ^B possible combinations of the information bits b ₀ , bi, b ₂ , b ₃ in a unique manner on 2 ^B combinations of the N 'output signals U ₀ , u _x , U ₂ , U ₃ , U ₄ from. Consequently, there exist 2 ^N '- 2 ^B combinations or signal combinations of the N' output signals U ₀ , Ui, U ₂ , U ₃ , U ₄ , which are not assigned to a bit pattern of the information bits bo, bi, b ₂ , b ₃ . Thus, while different (binary) signal combinations would be transferable via the N 'lines or channels 130_0, 130_l, 130_2, 130_3, 130_4 2 ^N ', less than the total possible 2 ^N 'signal combinations are actually used. In other words, a true subset of the 2 ^N 'possible signal combinations is used for transmission over the N' lines or channels.

There is thus a freedom in selecting the actual signal combinations used for the transmission of the B information bits over the N 'lines. The corresponding freedom makes it possible to use those signal combinations for transmission over the channel or lines 130, which, in particular taking into account the channel properties and the properties of the quantization, provide a sufficiently reliable receiver-side reconstruction or estimation of the information bits b ₀ , E ₁ , b ₂ , b ₃ allow.

It should be noted here that on the part of the signal decoder only such signal combinations of the transmission signals Uo, Ui, U ₂ , U ₃ , U ₄ lead to distinguishable reconstructed information bits B ₀ , E ₁ , b ₂ , S ₃ , for which a distinguishable Combination of the threshold-valued received signals zo, Zi, z ₂ , z ₃ , z ₄ results. Therefore, in the preferred embodiment, the vector imagers 140 are designed to generate, for different bit patterns of the information bits to be transmitted, only those signal combinations of the output signals u ₀ , Ui, U ₂ , u ₃ , u _{4 of} the signal coder 110 that produce distinguishable combinations of threshold values. evaluated receive signals Zo, Zi, Z ₂ , Z ₃ , Z ₄ lead. The corresponding selection of the signal combinations of the output signals uo, Ui, U ₂ , U ₃ , U ₄ can be determined, for example, metrologically or simulatively by more be used as 2 ^B signal combinations for the output signals U ₀ / Ui, U _2f U ₃ , U _{4 of} the signal encoder 110, and by to the various more than 2 ^B signal combinations of the output signals associated receive signals y ₀ , yi, yi, Y _{3 /} Y ₄ or threshold-valued received signals zo, Zi, Z ₂ , Z ₃ , Z _{4 are} determined. In the corresponding measurement or simulation, the transmission characteristic of the channel, for example, a crosstalk behavior of the channel, is taken into account. Of the more than 2 ^B used signal combinations of the output signals u ₀ , Ui, U ₂ , U ₃ , U ₄ then 2 ^B signal combinations are selected, the distinguishable signal combinations of the threshold-valued received signals Zo, Zi, Z ₂ , Z _3rd , Z ₄ , or lead to the best distinguishable combinations of the received signals. Subsequently, the vector imager 140 is configured (for example, by adjusting the mapping rule) to map the 2 ^B bit patterns of the B information bits to the selected signal combinations.

In addition to the requirement that the selected signal combination in the presence of a noise-free channel or noise-free transmission lead to distinguishable threshold-valued received signals Zo, Zi, z ₂ , z ₃ , z ₄ , can also be considered in the selection how reliable transmission, for example, in the presence of noise occurs. In other words, assuming that a particular signal combination is output at outputs uo, Ui, U ₂ , U ₃ , u _{4 of} a signal encoder 110, assuming a noise, probabilities can be determined with which the threshold decision 150 can determine different combinations of the threshold-weighted received signals z ₀ , zi, z _Σ , z ₃ , z ₄ outputs. Thus, computationally, simulatively or metrologically, it can be ascertained which signal combinations of the output signals of the signal coder 110 can be recognized and / or distinguished from one another at the receiving end, such as with high reliability. Accordingly, in a preferred embodiment, those signal combinations of the output signals of the signal coder are selected for use by the vector imager 140, the use of which results in at least one predetermined reliability of the threshold-valued received signals Zo, Z ₁ , Z ₂ , Z ₃ , z ₄ .

In another preferred embodiment, such signal combinations of the output signals of the signal coder are selected for use by the vector imager 140, which results in receive signals y ₀ , yi, y2, y ₃ , yi of the signal decoder 120 which are at least a predetermined minimum distance according to a predetermined standard exhibit.

In a further preferred embodiment, of a plurality of signal combinations at the output of the vector image 140, which lead to received signals (or received signal combinations) of the signal decoder whose distance is smaller than a predetermined minimum distance according to a predetermined standard, or the identical threshold value. evaluated receive signals Zo, Zi, Z ₂ , Z ₃ , Z ₄ , at most one signal combination selected for use by the vector encoder 140.

Incidentally, it should be noted that in selecting the signal combinations used by the vector imagers 140, the various criteria described above may also be combined with each other.

In summary, it can be stated that out of the total possible 2 ^N 'signal combinations on the output signals of the vector coder 110, those signal combinations are selected for use which, after being transmitted via the channel 130 to the signal decoder 120, lead to threshold-valued received signals allow a reliable reconstruction of the estimated information bits b ₀ , E ₁ , b ₂ , b ₃ . Instead of the highest possible reliability can at least on a reliability that is greater than a given minimum reliability is turned off.

In other words, in the transmission from the signal encoder 110 to the signal decoder 120, those signal combinations on the output signals of the signal encoder 110 are used, resulting in a bit error rate that is smaller than a predetermined bit error rate.

In other words, the vector imager 140 is designed not to form the combinations of the information bits bo, bi, b ₂ , b ₃ on signal combinations of the output signals of the signal coder 110 which have been evaluated as indistinguishable threshold-valued received signals or as threshold-valued Receive signals whose reliability is less than a predetermined minimum reliability lead.

The vector image 152 on the side of the signal decoder 120 is preferably designed to enable the most reliable reconstruction of the information bits bo, bi, b ₂ , i> ₃ based on the threshold-valued receive signals zo, Zi, Z ₂ , Z ₃ , Z ₄ to obtain estimated information bits B ₀ , B ₁ , B ₂ , B ₃ .

For this purpose, the vector mapper 152 of the signal decoder 120 is preferably configured to map combinations of thresholded received signals Z ₀ , Zi, z ₂ , Z ₃ , Z ₄ to estimated information bits B ₀ , B ₁ , B ₂ , B ₃ that at least in the presence of a noise-free channel 130 between the signal encoder 110 and the signal decoder 120, the estimated information bits B ₀ , B ₁ , B ₂ , B ₃ match the transmitted information bits bo, bi, b ₂ , b3. The configuration of the vector image 152 can be done, for example, by determining the associated combinations of the threshold-weighted received signals z ₀ , Zi, Z ₂ , Z ₃ , Z ₄ for the possible combinations of the information bits bo, bi, b ₂ , b ₃ , The vector image 142 is then configured so that the resulting combinations of the threshold-valued received signals are in turn mapped to associated reconstructed information bits b ₀ , S ₁ , S ₂ , b ₃ that match the transmitted information bits bo, bi, b ₂ , b ₃ , or which are associated with the transmitted information bits.

In this case, since the above-described configuration of the vector imager 152 of the signal decoder 120 is typically performed for an undistorted channel, only the mapping of 2 ^B combinations of the threshold-valued received signals from a total of 2 ^N 'possible combinations of threshold-valued received signals on combinations of B reconstructed information bits b ₀ , b _x , b ₂ , S ₃ known.

It is preferred, however, for the remaining 2 ^N> - 2 ^B

Combinations of the threshold-valued received signals

(which do not occur in an undisturbed case) define an image for associated combinations of the reconstructed information bits.

In order to thus define a vector map of the 2 ^{N '} - 2 ^B combinations of the threshold-valued received signals which do not occur in an undisturbed state, it can be determined, for example, in which combination of information bits to be sent bo, bi, b _Σ rb ₃ at the Input of the signal encoder 110 a contemplated combination of threshold-valued received signals in the presence of noise on the channel 130 occurs with a maximum probability. In other words, from the knowledge of the receive signals yor yi _/ Y 2 _f Y _3r Y4 occurring in a noise-free case at the input of the signal decoder 120 for different bit patterns of the information bits bo, bi, b ₂ , b _{3, it} can be determined with what probability the Bit pattern of the threshold-valued receive signals zo _/ Zi, Z ₂ , Z ₃ , Z ₄ for different combinations of information bits bo, bi, b ₂ , b ₃ occur. It can therefore For example, conditional probabilities are determined that describe the likelihood of having certain bit patterns present at the input of the signal encoder 110 for a contemplated combination of the thresholded received signals. The considered combination of the threshold-valued received signals may then in a preferred embodiment be associated with a combination of estimated information bits B ₀ , B ₁ , B ₂ , B ₃ which is identical to the transmitting-side combination of information bits, for which there is a maximum conditional probability consists.

In the aforementioned manner, for example, the mapping rule of the vector image 152 can be set to a minimum bit error rate in the transmission of the information bits b ₀ , bi, b ₂ / b ₃ or in the reconstruction of the reconstructed information bits B ₀ , B ₁ , B ₂ , B ₃ to achieve.

It should be noted, by the way, that the present invention is not limited to a certain number of simultaneously transmitted information bits or lines used. Rather, according to the basic idea of the present invention, it is only necessary that there are more lines N 'than information bits B are transmitted simultaneously (N'> B).

It should also be pointed out that the bus system 100 according to the invention according to FIG. 1 is based on the bus system 1800 according to FIG. 18 already described above. The bus system 100 according to the invention differs from the bus system 1800 according to FIG. 18, inter alia in that

1. the bus system according to the invention is expanded by an additional line;

2. at a transmitter (or in the signal encoder 110) a redundant room code is inserted; and 3. at a receiver (for example in the signal decoder 120) the redundant room code is decoded again.

The bus system 100 according to the invention according to FIG. 1 thus provides a remedy for the problem existing in the conventional bus system 1800 that a reliable transmission is not possible if the crosstalk between the individual lines is too pronounced.

Furthermore, a concrete example of a bus system according to the invention will be described below with reference to FIG. 1, in which four bits are transmitted on five lines using a redundant room code. Details with regard to an algorithm of the choice of the spatial code as well as with regard to a reliability of the bus system according to the invention are also explained in more detail below.

Fig. 2 shows a block diagram of a bus system according to the invention according to an embodiment of the present invention. The bus system according to FIG. 2 is designated in its entirety by 200. Since the bus system 200 according to FIG. 2 is very similar to the bus system 100 according to FIG. 1, the same devices and signals in the bus systems 100 and 200 are designated by the same reference numerals and will not be explained again here.

It should be noted, however, that in the bus system 200, the transmission of the output signals of the signal encoder 110 to the inputs of the signal decoder 120 via a plurality of parallel lines 130_0, 130_l, 130_2, 130_3, 130_4. The lines 130_0, 130_l, 130_2, 130_3, 130_4 are arranged so close to one another that an inductive, capacitive and / or conductive coupling occurs between the lines. In a preferred embodiment, the lines 130_0, 130_l, 130_2, 130_3, 130_4 extend parallel to one another at least for part of their total length. In this case, the second line 130_1 preferably has a smaller distance to the first line 130_0 than to the fourth line 130_3 and to the fifth line 130_4. Furthermore, the second line 130_l preferably has a smaller distance to the third line 130_2 than to the fourth line 130_3 and to the fifth line 130_4. Consequently, the coupling between the second line and the first line is typically stronger than between the second line and the fourth line, and also stronger than between the second line 130_l and the fifth line 130_4. In addition, the coupling between the second line 130_l and the third line 130_2 is typically stronger than the coupling between the second line 130_l and the fourth line 130_3. In addition, typically, the coupling between the second line 130_l and the third line 130_2 is typically stronger than the coupling between the second line 130_l and the fifth line 130_4.

Further, typically the distance between the third line 130_2 and the second line 130_l is less than the distance between the third line 130_2 and the first line 130_0. The distance between the third line 130_2 and the fourth line 130_3 is also typically smaller than the distance between the third line 130_2 and the first line 13O_O, and further also smaller than the distance between the third line 130_2 and the fifth line 130_4. Thus, typically a coupling between the third line 130_2 and the second line 130_l is stronger than a coupling between the third line 130_2 and the first line 130_0, and also stronger than a coupling between the third line 130_2 and the fifth line 130_4. Furthermore, a coupling between the third line 130_2 and the fourth line 130_3 is typically typical. This may be stronger than a coupling between the third line 130_2 and the first line 130_0, and stronger than a coupling between the third line 130_2 and the fifth line 130_4.

A distance between the fourth line 130_3 and the third line 130_2 is typically smaller than a distance between the fourth line 130_3 and the second line 130_l, and also smaller than a distance between the fourth line 130_3 and the first line 130_0. Thus, typically, a coupling between the fourth line 130_3 and the third line 130_2 is stronger than a coupling between the fourth line 130_3 and the first line 130_0, and also stronger than a coupling between the fourth line 130_3 and the second line 130_l. A distance between the fourth line 130_3 and the fifth line 130_4 is typically smaller than a distance between the fourth line 130_3 and the second line 130_l, and also smaller than a distance between the fourth line 130_3 and the first line 130_0. As a result, coupling between the fourth line 130_3 and the fifth line 130_4 is stronger than coupling between the fourth line 130_3 and the second line 130_l, and also stronger than coupling between the fourth line 130_3 and the first line 130_0.

Further, typically, a distance between the fifth line 130_4 and the fourth line 130_3 is smaller than a distance between the fifth line 130_4 and the third line 130_2, smaller than a distance between the fifth line 130_4 and the second line 130_51, and also smaller than one Distance between the fifth line 130_4 and the first line 130_0. Thus, a coupling between the fifth line 130_4 and the fourth line 130_3 is stronger than a coupling between the fifth line 130_4 and the third line 130_2, stronger than a coupling between the fifth line 130_4 and the second line 13O-I, as well as stronger than a coupling between the fifth line 130_4 and the first line 130_0.

In another embodiment, it is preferred that the lines 130_0, 130_l, 130_2, 130_3, 130_4 are arranged such that the second line 130_l is between the first line 130_0 and the fourth line 130_3, that the third line 130_2 is between the second line 130_l and the fourth line 130_3, and that the fourth line 130_3 is between the third line 130_2 and the fifth line 130_4. Thus, therefore, couplings between immediately adjacent lines (ie between lines 130_i, 130_i + 1, where l <= i <= 4) are typically stronger than couplings between more distant lines. Further, typically, the first line 130_0 and the fifth line 130_4 have only one immediately adjacent line of the five lines 130_0, 130_l, 130__2, 130_3, 130_4, whereas the second line 130_1, the third line 130_2, and the fourth line 130_3 have their geometrical Arrangement two immediately adjacent lines.

It should be noted, moreover, that a coupling between two adjacent lines can be given for example due to an electrical conductivity or due to an inductive or capacitive coupling. Coupling is here understood to mean that a signal which is impressed on one of the five lines 130_0, 130_1, 130_2, 130_3, 130_4 at an associated output of the signal coder, in an electrical response at an input of the signal decoder 120 belonging to another line results. In other words, a signal is coupled from one line to another line, or a signal on one line influences the voltage at the receiver-side end of another line. With regard to the coupling, it is also assumed that the lines are such that a coupling between directly adjacent lines is stronger than a coupling between lines further away or between lines which are not immediately adjacent (between them, therefore still another line runs).

With regard to the lines 130_0, 130_l, 130_2, 130_3, 130_4, FIG. 3 shows a schematic representation of five lines which are arranged on a chip, a substrate or a circuit board. The arrangement according to FIG. 3 is designated in its entirety by 300. The arrangement 300 comprises five parallel lines 130_0, 130_l, 130_2, 130_3, 130_4, which are arranged on a substrate as parallel planar lines. The second line 130_l extends at least in a section between the first line 130_0 and the third line 130_2. The third line 130_2 runs at least in a section between the second line 130_1 and the fourth line 130_3. The fourth line 130_3 extends at least in a section between the third line 130_2 and the fifth line 130_4. This results in a coupling between the lines, as has already been described between.

The lines may be, for example, lines that run exclusively on a chip. However, the lines between the signal encoder 110 and the signal decoder 120 can also run on a printed circuit or on a circuit board. Furthermore, the lines 130_0, 130_l, 130_2, 130_3, 130_4 can connect a plurality of chips and thereby comprise, at least as sections, bonding wires, plated-through holes, curves and / or creases. Particular advantages of the present invention arise, for example, when the lines are arranged so that a coupling between the lines described above occurs. With regard to the circuit arrangement 200 according to FIG. 2, it is further to be noted that the circuit arrangement 200 optionally comprises a transmission vector image adjuster 210 and / or a reception vector image adjuster 220. The optional transmit vector image adjuster 210 is designed to receive information about a channel, ie for example information about the transmission properties of the lines 130_0, 130_l, 130_2, 130_3, 130_4, and / or via a coupling between the lines. Based on the information, via the channel, the vector image adjusts the mapping scheme of the transmit vector mapper 140. The transmission vector image adjuster 210 can use one of the algorithms already described above or which will be described below. The aim of setting the mapping rule of the transmit vector mapper 140 is typically to minimize the bit error rate in the transmission of information bits bo, bi, b ₂ , b ₃ from the signal encoder 110 to the signal decoder 120 and in the reconstruction of the estimated information bits B. ₀ , S ₁ , S ₂ , b ₃ to achieve.

Optionally, the communication system 200 further includes a reception vector mapper 220 configured to set an imaging policy of the reception vector map 152 in response to information about the channel. One of the above-described algorithms, or alternatively one of the algorithms described below, may be used to adjust the mapping policy of the receive vector map 152.

It should be noted, moreover, that the adjustment of the mapping rule of the transmit vector mapper 140 and the receive vector mapper 152 can be done either by a one-time fixed programming of the transmit vector mapper 140 or several times by a repeated one Reprogramming the transmit vector mapper 140. Similarly, adjustment of the mapping policy of the receive vector mapper may be accomplished by a one time fixed program or by a multiple reprogramming.

Figure 4a shows a schematic representation of a memory for use as a receive vector imager 152. The memory of Figure 4a is designated 400 in its entirety. Memory 400 includes a plurality of address inputs 410 and a plurality of data outputs 420. Address inputs 410 are configured to receive a plurality of threshold-valued receive signals z ₀ , Zi, Z ₂ , Z ₃ , Z ₄ and are for selection In other words, based on a combination of values of the threshold-valued received signals at the address inputs 410 of the memory 400, the memory 400 selects one of its data fields. Typically, the address inputs are binary inputs, each of which may have two states. In the transmission system with N 'lines or channels, the memory 400 typically has N' address inputs, so that a total of 2 ^{N '} data fields are selectable. The memory 400 is, moreover, designed to output the information stored in the data field selected via the address inputs 410 at the data outputs 420. The data at the data outputs 420 thus serve as estimated or reconstructed information bits S ₀ , S ₁ , S ₂ , S ₃ . In a transmission system in which B bits are transmitted simultaneously over the N 'lines or channels, the memory 400 preferably has B parallel outputs. Thus, the memory 400 serves as a vector map which maps combinations of the N 'threshold-valued received signals to combinations of B reconstructed information bits S ₀ , S ₁ , S ₂ , S ₃ .

Incidentally, the memory 400 can be designed to permit a change in the data contents of the memory fields. chen. In other words, the memory 400 may be a random access memory. In this case, the mapping rule included in the memory 400 is set by the reception vector mapper 220, for example. Alternatively, a memory 400 may also be a read-only memory or a once-programmable memory. In this case, the mapping rule for mapping the signal combinations of the threshold-valued received signals to the reconstructed information bits is determined by a one-time and no longer changeable programming.

Fig. 4b shows a schematic representation of a memory for use as a transmit vector imager. The memory according to FIG. 4b is designated in its entirety by 450. The memory 450 comprises a plurality of address inputs 460 and a plurality of data outputs 470. The memory 450 is designed to receive a plurality of information bits bo, bi, b ₂ , b ₃ (preferably in parallel) via the address inputs. The memory 450 comprises a plurality of data fields which can be selected by the address inputs or by the signal combination applied to the address inputs. The memory 450 is further configured to provide the data content of a selected data field at the data outputs 470. The data outputs 470 thereby define a signal combination on a plurality of N 'lines, the data outputs 470 driving the lines either directly or via a line driver 142.

The address inputs 460 of the memory 450 are typically inputs for binary signals. If the memory B has address inputs to which the B information bits can be applied in parallel, the memory 450 typically comprises 2 ^B data fields, so that each combination of information bits is assigned exactly one data field. The information contained in the data field associated with a combination of the information bits is then passed over the Data outputs 470 output. The memory 450 thus assigns to each possible combination of information bits b ₀ , bi, b ₂ , b ₃ exactly one signal combination Co, Ci, C ₂ , C ₃ , C ₄ on the N 'parallel lines.

The memory 450 according to FIG. 4B may, for example, be a read-write memory. In this case, the mapping rule defined by the content of the memory fields of the memory 450 may be changed. The setting of the mapping rule can for example be done via the transmit vector image adjuster 210. Alternatively, however, the memory 450 may also be a once-programmable or mask-programmed memory, so that the mapping rule can not be changed after a single determination.

FIG. 5 shows a tabular representation of an association between bit combinations to be transmitted (b ₃ , b ₂ , bi, bo) and signal combinations (C ₄ , C ₃ , C ₂ , S ₂ , B ₂ ) to be transmitted on the transmission lines 130_ 0, 130_ 1, 130_ 2, 130_ 3, 130_ ₄ . Ci, Co). Furthermore, FIG. 5 shows a tabular description of a mapping rule between received, threshold-weighted received signals (z ₄ , Z ₃ , Z ₂ , Zi, Zo) and reconstructed or decoded information bits [B ₃ , S ₁ , S _x , S ₀ ). A first table or transmit table 510 according to FIG. 5 thus defines which signal combinations (c _4r C ₃ , C ₂ , C ₁ , C ₀ ) at the outputs of the vector mapper 140 correspond to the possible bit patterns (b ₃ , b ₂ , b _x , bo). at the entrances of the vector image 140 are assigned. Thus, the transmission table 510 defines which signal combinations (u4, U _3, U _2, U, Uo) are applied to the inputs of the lines 130_4, 130_3, 130_2, 130_l, 130_0 for different bit pattern of the information bits. It should be noted that, with respect to the transmit table 510, it is assumed that there are 2 ⁴ = 16 bit patterns to be transmitted (b3, b ₂ , bi, bo) which transmit over N '= 5 lines become. The 16 bit patterns of the information bits to be transmitted are plotted in a first column 512 of the transmit table 510. On the N '= 5 lines exist 2 ⁵ = 32 different signal combinations. The transmit table 510 defines 16 selected signal combinations that are applied to the inputs of the five lines 130_0, 130_l, 130_2, 130_3, 130_4 in dependence on the information bits to be transmitted. The selected signal combinations are applied to a second column 514 of the transmit table 510, bitmaps and signal combinations indicated in a same row in the first column 512 of the transmit table 510 and in the second column 514 being associated with each other. The remaining 16 possible signal combinations, which could still be applied to the inputs of the lines 130_0, 130_l, 130_2, 130_3, 130_4, but which are not entered in the second column 514 of the transmission table 510, are not used. The fact that according to the transmission table 510 thus only half of the total possible signal combinations on the lines 130 are used, corresponds to a spatial redundancy. Namely, according to the transmission table 510, only the signal combinations (Cή, C ₃ , C ₂ , Ci, Co) are assigned to possible bit patterns (b ₃ , t> 2 _r b, b ₀ ) of the information bits, on the receiver side, ie in the signal decoder 120, can be decoded with sufficient reliability, or from which the receiver side with sufficient reliability (eg with a bit error rate <10 ^{~ 3} , but preferably with a bit error rate <10 ^"4 or <10 ^{~ 8} ), the transmitted bit pattern (b ₃ , hz, bi, bo) can be reconstructed.

It should be noted, moreover, that it was assumed in the preparation of the transmission table 510 of a geometric arrangement of the lines, as described for example with reference to FIGS. 2 and 3. In other words, the transmission table is optimized for a certain predetermined or previously known line structure, in which between each electrically adjacent lines Strongest coupling (stronger than between more distant lines) or a strongest crosstalk occurs.

In other words, the transmission table 510 is based on an assignment of signals as well as an assignment of a layer of lines, as described with reference to FIGS. 1, 2 and 3.

FIG. 5 further shows a receive table 520 which is a mapping of threshold valued combinations of receive signals (z ₄ , Z ₃ , Z ₂ , Zi, zo) to combinations of decoded and reconstructed information _bits (b _3r b ₂ , b _lf b ₀ ) describes. The reception table 520 describes in a first column 522 combinations of threshold-valued received signals (z ₄ , Z ₃ , Z ₂ , Zi, zo). The receive table 520 shows in a second column 524 combinations of decoded and reconstructed or reconstructed information bits (b ₃ , S ₂ , S ₁ , b ₀ ). The first column 522 of

Receiving table 520 shows in an upper half 16 combinations of threshold-valued received signals (z ₄ , Z ₃ / Z ₂ »Zi / Zo) which, assuming a noise-free transmission, become transmitted signal combinations (c ₄ , C ₃ , C ₂ , Ci, Co) according to the second column 514 of the transmission table 510. The threshold-valued combinations of received signals (z ₄ , Z ₃ , Z ₂ , Zi, Zo) differ in part because of the crosstalk of the channel from the associated (arranged in a same row) transmitted signal combinations (c ₄ , C ₃ , C2 , Ci, Co). However, the transmission table 510 is selected such that the 16 different combinations of information bits to be transmitted (b ₃ , b ₂ , bi, bo) 16 different combinations of threshold-value received signals (z ₄ , Z ₃ , Z ₂ , Z ₁ , Zo) assigned. For this reason, at least assuming a noise-free transmission, an unambiguous assignment of the 16 combinations of threshold-valued received signals (z ₄ , Z ₃ , Z ₂ , Zi, Z ₀ ) shown in the upper half of column 522 is to be decoded or reconstructed - structured combinations of information bits (b ₃ , b ₂ , b _x , b ₀ ) possible.

Furthermore, there are 16 further combinations of the threshold-valued receive signals (z ₄ , z ₃ , zz, Zi, z ₀ ), which do not occur in an undisturbed or noise-free operation of the bus system 100. The further combinations of the threshold-valued receive signals (z ₄ , Z ₃ , Z ₂ , Zi, zo) are shown in the lower half of the first column 522 of the receive table 520. Assuming that the further combinations of the threshold-valued received signals (Z4, Z ₃ , Z2, Zi, Zo) do not occur, the further combinations could be given arbitrary combinations of decoded or reconstructed or estimated information bits (£ ₃ , S ₂ , b _x , b ₀ ).

However, it is preferred in a noise-free case on the receiving side not occurring further combinations of threshold-valued received signals (z ₄ , z ₃ , Z ₂ , Zi, Zo) such combinations of decoded information bits (6 ₃ , S ₂ , S ₁ , O ₀ ) at whose transmitting side presence the corresponding further combinations occur with a highest probability. In other words, for a considered combination of threshold-valued received signals (z ₄ , Z ₃ , z ₂ , Zi, z ₀ ), it can be calculated with what probability the combination considered in the presence of different transmit-side combinations of information bits (b ₃ , b ₂ , bi, bo) or in the presence of different transmit-side signal combinations (c ₄ , C ₃ , C ₂ , Ci, Co) occurs when a specific, assumed noise of, for example, the line driver 142, the channel 130 and / or or the Schwellwertentscheiders 150 is assumed. On the other hand, it can also be calculated by means of stochastics how large the probability is that a particular bit pattern of the information patterns (b ₃ , b ₂ , h _\ , bo) was sent if the considered combination of threshold values received receive signals (z ₄ , Z ₃ , 7.2, Zi, Zo) is received (conditional probability). Thus, in a preferred embodiment, the vector image of the receive vector mapper 152 is adjusted so that a considered (further) combination of threshold valued received signals (Z ₄ , z ₃ , Z ₂ , Zi, z <j) is limited to such a combination of decoded and / or decoded input signals reconstructed information bits (S ₃ , S ₂ , S ₁ , S ₀ ), which was transmitted by the signal coder 110 in the presence of the considered combination of threshold-valued received signals with the highest probability.

Thus, the invention redundant spatial coding considerable freedom in the choice of the receiving side vector image of the vector image 152 of the signal decoder 120. The signal decoder 120 is capable of more states or signal combinations of the received signals (y ₄ , V ₃ , γ ₂ , yi, yo) or the threshold-valued received signals (Z ₄ , Z ₃ , Z ₂ , Zi, Z ₀ ) to be distinguished as actually occur in a trouble-free or a (noise-free) transmission. The additional combinations, which do not occur in a noise-free transmission, can be mapped to decoded or reconstructed combinations of information bits (S ₃ , S ₂ , S ₁ , S ₀ ) by a suitable vector image such that a maximum

Reliability of the decoded or reconstructed information bits (b ₃ , S ₂ , S _x , b ₀ ) even in the presence of

Noise (for example noise) during transmission can be guaranteed.

The reception table 520 shown with reference to FIG. 5 describes a particularly advantageous assignment of threshold-value-received signals (z ₄ , Z ₃ , Z ₂ , Z ₁ , zo) to decoded or reconstructed combinations of information bits (S ₃ , S ₂ , S ₁ , S ₀ ). The mapping rule for the vector image 152 of the signal decoder 120 described by the reception table 520 is particularly advantageous in FIG Combination with an arrangement of the lines 130_0, 130_l, 130_2, 130_3, 130_4, as described for example with reference to FIGS. 1, 2 and 3.

In the following, the principle of redundant space coding will be described again generally with reference to FIG. 6. Fig. 6 shows a schematic representation of a digital transmission system. The transmission system according to FIG. 6 is designated in its entirety by 600. An (optional) serial / parallel converter 620 receives a bit stream 610, for example from a channel coding. The bit stream 610 may be, for example, a serial bit stream which, for example, within one time unit or within a predetermined time interval comprises B bits (Ci, C ₂ ,..., C _B ) (also referred to as information bits to be transmitted). A serial / parallel converter 620 receives the bit stream 610 and converts it into N sets 622_1, 622_2, ..., 622_N of one or more bits. An i-th set 622_i comprises b ± bits. The circuit arrangement 600 furthermore comprises N individual signal images 626_1,

626_2, 626_N. An i-th individual signal image 626_i receives the ith set 622_i of bi bits and maps the bi bits to a single modulation symbol X ₁ . In other words, the i-th single modulation symbol Xi is based on the bi bits of the ith set 622_i of bits. A single modulation symbol Xi can typically assume 2 'different values. By the way, the single modulation symbol Xi can either assume different real values or one or more different complex values. Incidentally, it should be noted that the single modulation symbol Xi is exclusively dependent on the bi bits of the ith set 622_i of information bits, but not on the remaining B-bi information bits. Namely, the individual signal images 626_i are designed to independently generate the respective single modulation symbols Xi. During which the circuit arrangement 600 B bits are supplied to a combination of N individual modulation symbols Xi, X2, • .., X _N • The various possible combinations of the individual modulation symbols Xi, X _2f ••• _/ X _N derive thereby a is thus produced per unit of time, Modulation alphabet M. In other words:

The modulation alphabet M typically comprises 2 ^B different combinations of single modulation symbols uniquely assigned by the individual signal images 626_i to the B bits (or the 2 ^B combinations of information bits representable by the B bits).

The combinations of single modulation symbols of Xi, X ₂ , ..., x _N are transmitted over an effective MIMO channel 630, as already described. For example, the effective MIMO channel 630 may include a modulator that receives various single modulation symbols xi, X2,, X _N within a time interval N and generates a modulated signal based thereon. At the outputs of the effective MIMO channel 630, for example, M are based on various individual received signals yi, y _Σ ,..., Y _M. The individual received signals yi, y _Σ, ..., y _M is typically analog signals. For this reason, the circuit arrangement 600 comprises a bank 640 of analog / digital converters 640_1, 640_2,..., 640_M. An i-th analog-to-digital converter 640_i receives the i-th receive signal y ± and generates an i-th quantized receive value Zj based thereon. The i-th quantized receive value z ± comprises ri bits. Incidentally, the M analog-to-digital converters 640_l, 640_2,, 640_M essentially operate independently of one another (ie with regard to the processing of received signals yi). Each of the M quantized reception values z ± is thus assigned to exactly one reception value yi. 6B shows a block diagram of a digital transmission system according to the invention using digital modulation and demodulation, according to an embodiment of the present invention. The circuit arrangement according to FIG. 6B is designated in its entirety by 650. The data transmission system 650 according to FIG. 6b is designed to receive a bit stream 660, for example, from the channel coding. Per unit of time or per channel use, an (optional) serial / parallel converter 670 receives B 'bits or information bits to be transmitted (ci, C2, ..., c _B ') - The serial / parallel converter thus gives per unit time per Channel use the B 'bits in a parallel form as bits Ci, C2 _/ ...., c _B - off. A vector signal space mapper 680 receives the B 'bits Ci, C ₂ , ...., C _{B' in} parallel and generates N single modulation symbols Xi, xΣr ...., XN-Der by a vectoring signal space map Vector signal space mapper 680 provides the

Combining the single modulation symbols Xi, X ₂ ,, x 'to an input of the effective MIMO channel 630. The vector signal mapper 680 thus provides the combination of the single modulation symbols Xi, X2, ...., x _N, for example, to an input of a modulator designed is to give the information described by the single modulation symbols Xi, X ₂ , ..., X _N during a time interval or during a channel use (simultaneously) to a transmission channel between a transmitter and a receiver. In other words, the modulator is, for example (but not necessarily), configured to simultaneously transmit the single modulation symbols Xi, X2,, XN over a wireless or wireline channel. It should be noted that the serial / parallel converter 670 and the vector images 680 are thus part of a signal coder 664.

At an output of the effective MIMO channel 630 are M receive signals yi, y2,. • .., yM, which are quantized by a set 640 of analog-to-digital converters receive quantized received signals Zi, Z ₂ ,, z _M.

Analog-to-digital conversion by analog-to-digital converter set 640 is done in the same manner as described with reference to FIG. 6A.

An essential component of the digital transmission system 650 is the vector signal space mapper 680, as described in detail below. The vector signal space mapper 680 receives B 'bits, preferably in parallel form. The vector signal space mapper 680 generates from the B 'bits a combination of N-single modulation symbols. However, due to the mapping scheme of the vector signal space mapper 680, there is no one-to-one correspondence between a particular single modulation symbol xi and a group of bits of the total B 'information bits Ci, C ₂ , ..., C _B '.

For example, there exists at least one single-modulation symbol Xi, which depends on j bits from the set of bits Ci, C ₂ , c _{B '} , and which altogether assumes less than 2- * different values. In other words, from a single one of the single modulation symbols Xi, X ₂ ,..., XN, no unambiguous inference is possible to a bit on which the single modulation symbol depends.

Incidentally, the vector signal space mapper 680 is designed to store the 2 ^{B '} possible combinations of the information bits Ci,

C ₂ , C _B ', which are used per channel use or per time interval, to map to 2 ^B ' different combinations of the single modulation symbols Xi, X ₂ , ...., x _N. It is assumed that the 2 ^B 'combinations of the N single modulation symbols Xi, X ₂ , ..,., X _{N form} a modulation alphabet M'. In other words:

[xi X ₂ x _N ] ^τ e M '.

The modulation alphabet M ^f is typically a true subset of a total modulation alphabet M, all possible combinations of the N-Einzelmodulationssyτnbole X ₁ , * 2r • _' •, X _N includes. In other words, the total modulation alphabet comprises all combinations of the occurring values of the single modulation symbols.

Thus, irrespective of the transmitted data or the transmitted information bits di, d ₂ ,..., C _' , not all possible combinations of the N individual modulation symbols X ₁ , X _2,. •. , X _{N occur} when mapping the B 'information bits through the vector signal space mapper 680, there is redundant space coding. In other words, not all discrete combinations of single modulation symbols that would be available according to a complete modulation alphabet M are used. Thus, it is possible to use those combinations of single modulation symbols which lead to reliably distinguishable received signals after transmission over the effective MIMO channel 630, whereas those combinations of single modulation symbols leading to difficult or indistinguishable received signals, Not used.

The vector signal space mapper 680 can be realized, for example, by a memory or a look-up table, as described for example with reference to FIG. 4B. The memory 450 according to FIG. 4A can be modified, for example, in order to receive the bits C ₁ , cΣ,..., C _B 'in parallel at its address inputs 460. At the data outputs 470, the memory 450 may, for example, output the real-valued or complex-valued individual modulation symbols X ₁ , X ₂ ,..., X _N in coded form.

But the vector signal space mapper 680 can be implemented in other ways, as long as it is ensured that a corresponding vector or linking mapping of the information bits C _1, C _2, .. •, c _B 'to a set of single-modulation symbols X _1, X ₂ , .-., XN is guaranteed. The mode of operation of the circuit arrangement 650 according to FIG. 6A will be explained again below. An essential core idea of the present invention is the use of a so-called redundant space coding in a signal mapping within a block of the digital modulation (see Fig. 14). In the following, the basic approach and, subsequently, the redundant spatial coding will be explained.

As stated above, coarse analog-to-digital conversion results in indistinguishable or hard-to-distinguish transmission signals for the receiver, which virtually prevents the occurrence of reliable communication (see Fig. 16 and the explanations thereof). This problem is solved by the present invention as follows:

1. The modulation alphabet is restricted (compared to an original modulation alphabet) so that not all possible signal vectors (or combinations of single modulation symbols) may be transmitted. This limitation of the modulation alphabet creates a redundant space code. There are two different ways to create such a redundant room code. These are explained below.

2. A bank of signal space mappings (or of individual signal mappings 626_1, 626_2,..., 626_N) originally present (thus for example in a transmission system according to FIGS. 6A and 7A) is represented by a vector signal space map or a Vector signal space mapper 680).

3. If it is necessary to achieve the same data rate as in an original system (that is, for example, as in a system according to FIGS. 6A or 7A), either a) increases the code rate of the channel code or channel coder, or

b) increases the number of inputs and / or outputs of the effective channel.

In the following, the said principle will be explained in more detail.

For this purpose, changes will be made in the following which, according to an embodiment of the present invention, are made on the basis of an original digital modulation and demodulation transmission system according to FIG. 6A.

In other words, according to an embodiment of the present invention, changes outlined in a block of digital modulation are made with reference to Figs. 6A and 6B. Here, Fig. 6A shows an original digital modulation / demodulation, while Fig. 6B shows a modified system according to an embodiment of the present invention. Essentially, the digital modulator is changed.

A new system 650 for digital modulation / demodulation originates from the original system 600 by using a vector signal space map, which implements a redundant room code.

The modified system 650 according to an embodiment of the present invention differs from the original system 600 essentially in the following points:

1. A number B of bits from the channel coder (or channel coder), which are processed per channel use by the digital modulation, is reduced (starting from a value B) to a value B ': B → B '<B.

2. The previous N signal space maps or individual signal space maps 626_1, 626_2,..., 626_N (also denoted by sigi, sig ₂ ,..., Sig _N ) are replaced by a single vector signal space map. The vector signal mapping 680 forms B 'bits from 2 ^B' possible send vectors from a new modulation alphabet AT.

3. The new modulation alphabet AT is characterized in that it contains fewer elements of the same dimension than the original modulation alphabet Af. A technically preferred embodiment is to generate the new modulation alphabet Af 'from the original modulation alphabet M by omitting elements:

M → M ^r c M.

The vector signal space map 680 describes a redundant space code. A redundant space code would also arise if the dimension of the vectors is increased with the same number of elements (vectors or combinations of single modulation symbols). This possibility will be discussed in detail below.

First, however, it will first be described how the room code can be selected. The redundant room code is not unique, which means there are many ways to design this room code. A technically particularly meaningful or advantageous expression is to obtain Af 'from Af by omitting those elements which are the most difficult to distinguish from each other in the receiver (that is, for example, in a signal decoder). This can be done in such a way that each vector X ₁ e Af (ie to each combination of single modulation symbols Xi _/ X _{2 /} - .. _/ X _N ) the following number φ (or a corresponding value) is assigned:

where z = [z _\ , z ₂ , ... Z _M _ ^{T is} a vector of analog / digital-converted received signals. Further is

x (z) = argmaxPr [x | z]

the most probable sending vector given the quantized observation z.

In other words, for a given quantized

Λ observation z, a most probable transmission vector x (z) is determined. Subsequently, a probability is determined that the estimated most probable transmit vector

Λ each (z) coincides with the actual transmitted transmission vector x ±, provided that the transmission vector is transmitted x ±. The corresponding probability is denoted by φ (xi).

The symbol Pr [.] Stands for the probability of a random event. The numbers φ (Xi) can be concretely calculated by known methods of mathematical statistics with known effective MIMO channel (channel matrix H), known statistics of the receiver noise and known characteristics of the analog / digital converter. The set M 'is then obtained from the 2 ^{B <} elements of M which have the largest associated numbers φ (Xi). These transmit vectors are best distinguished after quantization at the receiver.

The following describes how, according to an exemplary embodiment of the present invention, the bit width or the bit width Number of information bits B 'transmitted per channel use is selected.

The smaller the bit width B 'chosen, the fewer elements M' contains, and the better these can be distinguished from each other after the coarse analog-to-digital conversion. However, this simultaneously increases the redundancy of the room code. This reduces the effective data rate. If the communication system (that is to say, for example, the original transmission system 600 according to FIG. 6A) uses a channel code of the code rate R ₀ , the incorporation of the redundant space code results in an effective code rate R _c , _ef f

In general, to make rate loss as small as possible, B '<B should only be made as small as necessary to achieve reliable communication despite the gross analog-to-digital conversion.

The following describes how the above rate loss can be compensated. In order to compensate for the rate loss due to the redundant space coding, there are in principle two possibilities:

1. Space coding with increased dimension of the modulation alphabet. This approach will be explained in more detail below.

2. Increase the code rate of the channel code. For example, the code rate of the channel code or channel coder can be changed from an existing code rate R _c to a new code rate R _c <we as follows:

This compensates for the rate loss produced by the transmission of only B 'bits per channel use or per unit of time (instead of originally B bits per channel usage or per unit of time). However, this is only possible if the original code rate R _c applies:

In the case of uncoded transmission (R _c = 1), the named method is therefore not applicable.

In the following, an example of how a transmission system according to the invention with a vector signal space map 680 from a conventional transmission system can arise.

Assume that in the original transmission system, the number of bits of information bits transmitted per channel use is 8, and that the code rate R _{c of} the original transmission system is equal to H. In formulas: B = 8 bits and R _c = ^. Given a choice of B '= 6 bits, ie if 6 information bits are transmitted per channel use in the transmission system according to the invention, the code rate of the channel code would have to be changed, for example by a Puncturing, also Rc = 2/3 increased to compensate for the rate loss.

The following describes a redundant space coding with a dimension extension. For this purpose, FIG. 7A shows a schematic representation of an original transmission system with digital modulation and demodulation. The circuit arrangement according to FIG. 7A is designated in its entirety by 700 and substantially corresponds to the circuit arrangement 600 according to FIG. 6A. Therefore, like devices and signals in Figs. 6A and 7A are denoted by like reference numerals and will not be explained again here.

FIG. 7B shows a schematic representation of a transmission system according to an exemplary embodiment of the present invention, which results from the circuit arrangement 700 according to FIG. 7A through the changes described below. Incidentally, the circuit arrangement according to FIG. 7B is designated in its entirety by 750.

The circuitry 750 receives a bitstream 760, for example, from the channel encoding. The bit stream 760 comprises per channel use or per unit time B 'bits Ci, C2, ..., c _B '. A serial / parallel converter 770 converts bitstream 760 of B 'bits into parallel bits Ci, C ₂ , ..., C _B - and provides them to a vector signal space mapper 780 as input signals in parallel form. The vector signal space mapper 780 is designed to provide N 'single modulation symbols Xi, X ₂ , ..., XN' during a time interval or during a channel usage. The combination of the individual modulation symbols Xi, x ₂ ,..., X _N 'forms a transmission vector x. An effective MIMO channel 784 receives the N 'single modulation symbols Xi, x ₂ , ..., x _N ' simultaneously, and generates received signals yi, Yz, ..., Yw based on the N 'single modulation symbols M'. From each of the M 'received signals is then through a corresponding analog / digital converter 790_l, 790_2, ..., 790_M a associated quantized received signal zi, z ₂ , ..., z _w generated.

The circuit arrangement 750 differs from the circuit arrangement essentially in that information bits are transmitted instead of B 'bits per time unit or per channel use, and that the vector signal space mapper 780 N' instead of N individual modulation symbols for transmission via the effective MIMO. Channel 784 are generated.

However, a basic functionality of the vector signal space mapper 780 does not differ from the basic functionality of the vector signal space mapper 680. Nor does the basic functionality of the MIMO effective channel 784 differ from the basic functionality of the MIMO effective channel 630.

Thus, according to one embodiment of the present invention, one way to compensate for the rate loss caused by redundant space coding is to increase the number of inputs and outputs of the effective MIMO channel 784 (over a conventionally used MIMO effective channel 630). This usually means that the number of inputs and outputs of the physical channel must be increased as well. In a radio transmission system, for example, further antennas must be installed at a transmitter and at a receiver, or additional frequency channels must be used. In a wired communication system, however, more lines must be used.

Such an increase in the number of inputs and outputs of the MIMO effective channel is systemically costly. Particularly expediently, said extension is, for example, in a wired bus system on a printed circuit board or within a integrated circuit, as this simply a few more lines are inserted. For example, a 16-bit wide bus can be run instead of 16 now with 17 lines. However, the application of the described concept in a bus system will be discussed in more detail below.

First, however, the changes that are made to the digital modulator / demodulator will be described in general terms. FIGS. 7A and 7B show the necessary changes to the digital modulator and demodulator. 7A shows an original digital modulation / demodulation, and FIG. 7B shows a modified system according to an exemplary embodiment of the present invention. The new system of digital modulation / demodulation according to FIG. 7B arises from the original system by increasing the number of inputs (N '> N) and outputs (M'> M) and by using a vector signal space map, the implemented a redundant room code.

More specifically, for variations of the system 750 of FIG. 7B relative to the system 700 of FIG. 7A:

1. The number of inputs of the effective channel 784 is increased (compared to the effective channel 630):

N → N '> N.

2. The number of effective channel 784 outputs is increased (versus effective channel 630):

M → M ^y > M.

3. The number B of bits taken per channel use (or per unit of time) from the output stream of the channel encoder or channel coder (and the when using a channel simultaneously over the effective MIMO channel 784) remains the same (as compared to a number of concurrently transmitted bits in the transmission system 600):

B → B '= B.

4. The previous N signal space mappings or individual signal mappings 626_1, 626_2,..., 626_N (also referred to as sigi, sig ₂ ,..., Sig _N ) are replaced by a vector signal space map 780. The vector signal space map 784 thereby maps B bits to 2 ^B possible transmit vectors x '(consisting of N' single modulation symbols x _x , xΣ,..., X _N ') from the new modulation alphabet M'. In other words, it holds:

[xi X2 ... XN '] ^τ e Λf.

5. The new modulation alphabet W is characterized by the fact that it contains just as many elements (vectors x or combinations of single modulation symbols) as the original modulation alphabet M. However, the elements (vectors x or combinations of single modulation symbols) have the dimension N ', which is greater than the dimension N of the vectors of M.

The following describes details regarding a choice of the room code. The redundant space code (ie the choice of the elements of the new modulation alphabet M ') is not unique, that is, there are numerous ways to make this room code. Essential to the present invention is merely that

1. the dimension of the elements of the (new) modulation alphabet M 'is greater than the dimension of the elements the modulation alphabet M of the original system; and

2. the number of elements of the modulation alphabet M 'corresponds to the number of elements of the modulation alphabet M.

This creates a redundant room code. It should be understood, however, that the present invention is not limited to the use of a particular redundant space code.

A technically particularly meaningful and advantageous expression of a redundant space code arises from the fact that the modulation alphabet M 'is determined as follows:

1. Set ΔN = N '- N

2. Define quantities MN ₊ I, M + 2r ••• _> -MN + ΔNΛ which consist of at least two elements. The present invention is not limited to any specific choice of elements. However, for example, sets of single modulation symbols can be used _H, Λ ^ ₊ A, for example, correspond to a 4 QAM modulation or 16QAM modulation - for the quantity ΛÜ _N. Alternatively, the elements of said sets may also include modulation values of BPSK modulation, QPSK modulation or similar modulation having at least two possible real or complex values for the single modulation symbols.

For example, if all the signal space maps 626_1, 626_2,..., 626_N are identical in the original system (as shown in FIG. 7A), it is useful to select the sets or individual modulation alphabets M ± in the following manner: M _x = Mi-1 for i € {2, 3,. , , , N + ΔN}.

Otherwise, the elements of the sets or individual modulation alphabets M _mi for {2,..., ΔN} can be selected according to other criteria.

3. Set M = M ₁ × M2 × ... x MN + Δ _N (where M describes a total modulation alphabet comprising all possible combinations of the single modulation symbols from the single modulation alphabets Mi).

4. Order each vector Xj. e M the following number:

Here z = [zi, Z ₂ ,..., Z _M '] ^{τ is} the vector of the analog / digital converted received signal (compare FIG.

7B). Further, x (z) is the most probable transmit vector given the quantized observation z, according to

Λ x (z) = arg max Pr [x | z]. xeΛ /

With regard to more detailed explanations, such as the

Λ estimated vector x (z) is determined _{from the} observation z, and how the value Φ is determined, _W i _rC i _au f referenced above.

5. Constructed the modulation alphabet M 'from the 2 ^B elements of M, which have the largest associated numbers φ (xi).

In other words, out of the total modulation alphabet M, those transmission vectors x or combinations of Single modulation signals selected which result in quantized received signals z, which allow the most reliable reconstruction of the transmitted bit patterns. The vector signal space mapper 780 is then configured to map the combinations of the B information bits Ci, C ₂ ,..., C _B to the 2 ^B elements of the modulation alphabet W.

8 further shows a schematic representation of a digital transmission system according to an embodiment of the present invention. The circuit arrangement according to FIG. 8 is designated in its entirety by 800. The circuitry 800 includes a signal encoder 810 and a signal decoder 820. The signal encoder 810 receives B 'information bits Ci, C ₂ , ..., c _B > in a parallel manner and generates a timing signal 830 based thereon. The signal encoder 810 includes a vector mapper 840 corresponding to the previously described vector images 680, 780. The vector imager 840 provides N single modulation symbols Xi, X ₂ , • based on the B 'information bits. •, X _{H /} which are represented, for example, in the form of real or complex numerical values. A modulator 850 receives the individual modulation symbols xi, _X2, ..., X _N in parallel form and generated based thereon, the time signal 830. The individual modulation symbols Xi, x _2, ..., x _N, for example, an in-phase component and describe a quadrature portion as used by modulator 850 in the generation of complex-valued in-phase quadrature modulation (IQ modulation). The modulator 850 may be, for example, a multichannel in-phase quadrature modulator. For example, the modulator may be configured 850 to transmit the first single modulation symbol Xi at a first carrier frequency, to transmit the second single modulation symbol X ₂ at a second carrier frequency and to transfer the n-th single modulation symbol in an n-th carrier frequency. The signal encoder 810 further optionally additionally includes a transmit vector mapper 860 configured to adjust the mapping scheme of the vector imager 840 based on information about the channel to ensure that the B 'information bits are transmitted with sufficient reliability.

The adjustment of the map of the vector image 840 is done by the optional transmit vector map adjuster 860, for example according to one of the algorithms for selecting the modulation alphabet described above.

In the time signal 830, the B 'information bits are thus simultaneously transmitted within the scope of a channel usage or within a time interval.

The signal decoder 820 includes a demodulator 860 that receives the time signal 830 after transmission over a channel (possibly after a frequency translation). The demodulator 870 is designed to generate, based on the time signal, a plurality of N receive signals yi, y2, ..-, Y _N 'ZU, the portions of the received time signal 830 in different channels (eg frequency channels, code channels). describe. In the receive signal yi, y2, ..., Y _N may be, for example, real-valued or complex-valued signals. For example, the demodulator 870 may be an in-phase quadrature demodulator configured to extract complex-valued modulation amplitudes from different channels or frequency channels of the timing signal 830.

The complex-valued modulation amplitudes thus form the received signals yi, Y _{2 /} - « _* Y _N. The signal decoder 620 further includes a block 880 of analog-to-digital converters or threshold decision makers. An i-th analog / digital converter 880_i receives the ith receive signal yi from the demodulator 870 and provides a quantized or Threshold-Evaluated Receiving Signal Zj. A vector imager 890 receives the quantized or thresholded received signals zi, Z ₂ ,..., Z _N and forms them according to a mapping rule on B '.

Λ Λ Λ reconstructs information bits c \, Ci and CB <. The mapping rule of the vector image 890 is set in order to minimize the bit error rate of the reconstructed image.

Informations Λ Λ constructed information bits c \, Ci and CB 'TO achieve.

The signal decoder 820 further optionally includes a receive vector image adjuster 894 configured to adjust the mapping law of the vector imager 890 based on information about the channel between the signal encoder 810 and the signal decoder 820. The information about the channel can be determined, for example, by a channel estimator, not shown here, as it is known from the field of mobile telephony. By dynamically adjusting the mapping scheme of the vector imager 890 in response to information about the channel, it can be achieved that the signal decoder is always operating at a minimum possible bit error rate.

In the following, the operation of the detector (also referred to as signal decoder or signal receiver) or changes of the detector compared to conventional detectors will be described. The task of the detector is to obtain estimated values for the channel-coded bits from the transmitter (or for the channel-coded information bits sent) from a quantized observation vector z. For this purpose, the detector must be aware of the modulation alphabet (used by the transmitter or signal coder). The procedure of the detection can be chosen differently, as long as the detector can calculate estimated values for the transmitted bits from the knowledge of the new modulation alphabet M 'and from the observation or from the observation vector z. This can be done, for example, with the aid of a MAP detector (or a maximum-a-posteriori-probability Detector or a maximum a posteriori probability detector). However, not every known detector can be used. In an original system, the modulation alphabet M can be represented as a Cartesian product of quantities Mi, M ₂ ,..., M $ of lesser thickness. Therefore, it is possible to perform the detection successively based on the sets (or single modulation alphabets) M ₁ , M ₂ ,..., Λfo. In contrast, the space-code-generating modulation alphabet M 'generally can not be represented as a Cartesian product of lesser thickness quantities. Accordingly, the detection must operate directly on the set M ^r . Thus, if a successive detection was performed in an original system, then the new system must be switched to a common detection based on the set M '. On the other hand, if the detection is already based on the set M in an original system, then no structural changes are necessary. Then only M has to be replaced by M ^r .

In other words, an inventive detector according to an embodiment of the present invention differs from a conventional detector in that the detector according to the invention comprises a vector image, the plurality of quantized received signals Zi, Z ₂ , ..., z _{N '} linked to estimates for the Maps information bits.

In the following, the core idea of the present invention will be briefly summarized again.

The essence of the present invention is that a vector signal space mapping is used which generates a modulation alphabet M 'containing either fewer elements of the same dimension or the same number of elements of higher dimension than an original modulation alphabet M. In both cases, a redundant room code. The purpose of the above-mentioned procedure se is to send only those symbols, which can be decoded even after an analog / digital conversion. The essential advantage of the inventive concept over the current state of the art is that it enables digital communication via coarsely-quantized multichannel systems, which does not work or does not work with sufficient quality without the use of the present invention. Since the room code has no temporal component, there is no additional delay time (latency) during detection (decoding). This is a particular advantage of the invention since it is also possible to use it, especially in extremely latency-sensitive applications (eg in the case of bus systems on or between integrated circuits).

In other words, redundancy is provided within a combination of single modulation symbols transmitted simultaneously across the channel. Both signal coding and signal decoding take into account the transmission characteristics of the channel and, in particular, crosstalk between individual modulation symbols or individual transmission lines.

An exemplary functional proof of the present invention will be described below. In order to show the advantages of the present invention, a bit error rate of an example system was determined with and without the use of the inventive concept.

As the original system or comparison system, the system according to FIG. 15 is used, wherein the channel matrix H of the effective channel is given as follows:

The effective channel or effective MIMO channel 1360 of the comparison system has N = 4 inputs and M = 4 outputs. The signal space constellation of the originating system or comparison system (which is generated by the individual signal images 1520_l, 1520_2, 1520_3, and 1520_4) is the popular 4QAM in practice. Accordingly, for the single modulation alphabets Mi, MΣ, M3, M4 '

This leads to a modulation alphabet or overall modulation alphabet M with 4 ⁴ = 256 elements. Furthermore, for the real part and the imaginary part of each of the M output signals, analog / digital converters with one bit resolution (ie, sign decoder) are assumed. The fact of a coarse quantization (analog / digital conversion) is given here. After the quantization, there are indistinguishable transmission vectors.

For comparison, a digital transmission system according to the present invention is further considered. In the course of using the present invention, the number of inputs and outputs of the effective channel is increased by one each. For the number N 'of the inputs of the effective channel and for the number M' of the outputs of the effective channel, then:

M '= M + 1 = 5, UAK / N' = N + 1 = 5

For the new channel matrix H '(using the inventive concept) it is assumed: - 6 U5j

Since Mi = M ₂ = M ₃ = M ₄ = {± 1 ± j} / V2 is set for M ₅ :

From the 1024 elements of the set M with M = Mi x M ₂ x M ₃ x M ₄ x M ₅

For example, 256 elements are selected according to one of the modulation alphabet determination methods described above to form the new modulation alphabet M ^r (where M represents the overall modulation alphabet).

As a channel coding method, a Viterbi-decoded convolutional code (the memory length 6) is adopted, and as a reception signal processing, a so-called log-likelihood detection is adopted.

Fig. 9 is a graph showing a bit error rate after Viterbi decoding of a convolutional code. The graph of FIG. 9 is designated 900 in its entirety. At an abscissa 910, a transmission power to noise power ratio in decibels is plotted. An ordinate 912 describes an encoded bit error rate in a range between 10 ^"4 and 10 ^~ °. A first curve 920 describes a dependence of the coded bit error rate of the transmission power-to-Rauschleistungs- ratio using a conventional arrangement with an analog / digital converter with arbitrarily high resolution A second curve 930 shows a dependence of the encoded bit error rate from the transmit power to noise power ratio for a conventional transmission system using a single bit analog to digital converter. A third plot 940 depicts a dependence of the coded bit error rate on the transmit power to noise power ratio for a transmission system according to the invention using a one bit analog to digital converter and a space code.

In other words, Fig. 9 shows the resulting bit error frequencies as a function of the ratio of applied transmit power P _τ to the power σ ² _{n of} the receive noise. Without use of the method of the invention or the inventive concept results (when using a one-bit analog / digital converter) a to a saturation value of 10 ^{~ 4} hinlaufende bit error rate (bit error rate identical to that of FIG. 17). By contrast, using the method according to the invention or the inventive concept, the system works much better. In particular, it does not come to the formation of a saturation of the bit error rate. The bit error frequencies decrease monotonically with increasing error performance. This demonstrates the functionality of the invention.

Hereinafter, still referring to Figs. 1 to 5, a bus system as an application example of the present invention will be described. A wired bus system for connecting integrated circuits on a circuit board or for connecting subsystems within an integrated circuit represents a typical technical realization of a digital communication via a 1-bit quantized multi-channel system. Namely, at a receiving end of the bus system are received signals binary decisions are made, that is, the received signals (yo, yi, yz _t Υ3 _t YA) with 1-bit A / D converters (or with threshold deciders 150_0, 150_l, 150_2, 150 3, 150 4 ) is quantized. By a physical proximity of individual lines of the bus system to each other arise electrical couplings, which lead to a crosstalk of the signals. This behavior limits the usable clock frequency of the bus system, since otherwise (for example, in a further increase in the clock frequency of the bus system), the crosstalk leads to a coarse quantization (ie, for example, to indistinguishable transmission vectors), which prevents the occurrence of successful communication in the current state of the art , The present invention can be used to successfully operate the system despite coarse quantization. The following is an example of how the present invention can be applied to a bus system.

It should be noted that an original bus system used for comparison in which 4 bits are transmitted on 4 lines has already been described with reference to FIGS. 18 and 19. The inventive bus system according to an embodiment of the present invention shown in FIG. 1 improves the known bus system by providing

1. the bus system is extended by an additional line;

2. a redundant room code is inserted at the transmitter; and

3. the receiver decodes the redundant room code again.

The correspondingly modified bus system is shown in FIG. 1. The redundant room code is used according to one of the methods already described above for selecting the redundant room code or for determining the image preview. font of the vector image selected. Here, the total modulation alphabet M is set:

M = {0 volts, 5 volts} ⁵

From the | For MI = 32 elements, exactly 16 elements are selected according to items 4 and 5 of the above-mentioned algorithm for determining the modulation alphabet M ^f . As a signal-to-noise ratio, a relatively high value of 50 decibels is assumed, since in a bus system the receiver noise is typically almost negligible. The selected 16 elements form the new modulation alphabet Af 'which defines the redundant space code. The resulting mapping rule is from the two columns 512, 514 of the transmit table 510 of FIG.

5 can be seen. The two columns 510, 512 of the table of

Fig. 5 shows a map of four bits to be transmitted

(b ₃ b ₂ bi bo) to the actually transmitted five bits (c ₄ C ₃

C2 Ci Co), which are then transmitted via the bus system extended by one line (to a total of five lines).

If, for example, a bit pattern (b ₃ b ₂ bi bo) = (1011) is to be transmitted, then the redundant room code maps this to a signal combination (C ₄ C ₃ C ₂ Ci Co) = (01010). The signal combination (01010) is then sent via the five bus lines 130_0, 130_l, 130_2, 130_3, 130_4.

The couplings of the lines are due to the effective MIMO channel matrix H 'according to

1 0.35 0.1225 0.0429 0.015

0.35 1 0.35 0.1225 0.0429

H '= 0.1225 0.35 1 0.35 0.1225

0.0429 0.1225 0.35 1 0.35

0.015 0.0429 0.1225 0.35 certainly. The effective MIMO channel matrix H 'arises from the original effective channel matrix by adding a row and a column. The effective MIMO channel matrix H 'describes the relationship between received signals y ₀ , Yi / Yi, Y3, Yn and transmission signals u ₀ , Ui, uΣ _f U ₃ , U ₄ . The following applies:

y = H u.

After a transmission over the five lines 130_0, 130__l, 130_2, 130_3, 130_4 of the new bus system, the receive voltages (yo, yi, yΣ, Y3r Yi) are as before (for example, using the threshold decision 150_0, 150_l, 150_2, 150_3, 150_4 ) are quantized to binary values (z ₀ , Z ₁ , z ₂ , Z ₃ , Z ₄ ) and fed to inputs of the detector 152 (also referred to as vector images). The detector or vector image 152 has the task, at his

Output to output estimates (Z> ₀ , 6,, ö ₂ , 6 ₃ ) for the bits or information bits (bo _/ bi, b ₂ , b ₃ ). This is done by mapping the 32 possible input values (or the 32 possible combinations of the threshold-valued received signals (zo, Zi, Z ₂ , Z ₃ , z ₄ ) to 16 output values (or combinations of the estimated information bits

(ό _o , £, z> ₂ , £ ₃ ). The two right-hand columns 522, 524 of the table according to FIG. 5 show these mappings and thus define the operations of the detector or vector imager on the receiver in the signal decoder 120. Without receiver-side noise, only the upper 16 entries of the receive table 520 of FIG Meaning, since the lower 16 received values or combinations of threshold-valued received signals (z ₄ Z ₃ Z ₂ Zi Z ₀ ) can only arise due to the influence of noise. Since wired bus systems operate with a very high signal-to-noise ratio (50 dB in the example), the influence of the noise is practically negligible. Therefore, the lower 16 rows of the receive table 520 need not necessarily be implemented. As an example, assume that the bit pattern (b ₃ ϊ> ₂ bi bo) = 1011 is transmitted. The redundant space code (of the signal coder) maps this (c ₄ C ₃ C ₂ C ₁ C ₀ ) = (01010) (see column 12 of the table according to FIG. The signal combination (c ₄ C ₃ C ₂ Ci Co) is received without noise as (z ₄ Z ₃ Z ₂ Zi Z ₀ ) = (OHIO). According to the two right columns 522, 524 of the table of Fig. 5, this reception value is mapped to estimated information bits {b _Qi b _x , b ₂ , b ₃ ) = (1011). Thus, the combination of estimated information bits corresponds exactly to the bit combination to be transmitted.

Similar to the example shown, each of the 16 possible bit combinations is such that reliable communication is achieved. The key to this lies in the use of the redundant space code in the transmitter or signal encoder, which transmits the four (bits to be transmitted) or bits of information (bo bi b ₂ b ₃ > over five lines.

Both the redundant space code at the transmitter (or the vector images of the signal coder) and the detector or vector images at the receiver or signal decoder can be implemented, for example, by a respective memory module which implements the transmit table or the receive table (see FIGS Fig. 4B). This creates a practical latency-free implementation, as required for a bus system.

It should be noted, moreover, that in the example shown, 80 bits of memory capacity are required for storing the transmission table (or the redundant space code or the mapping rule of the vector image in the signal encoder). Furthermore, 128 bits of memory capacity are used to store the receive table in the detector (or to store the image). writing the vector image in the signal decoder).

10A shows a flow chart of a method of configuring a signal encoder for generating N 'parallel binary signals based on B information bits. It is assumed here that the signal coder essentially corresponds in terms of its structure to the signal coder 110, that is to say the signal coder has, for example, a vector image 140 for non-linear mapping of possible bit patterns of the B information bits (bo bi b ₂ b ₃ ) to 2 ^B signal combinations on the N 'parallel binary signals (uo, Ui, U ₂ , U ₃ , U ₄ ) according to a mapping rule, and further that: N'> B. The method according to the Fig. 10A is in its entirety with 1000 designates. The method 1000 comprises in a first step 1010 a determination of received signals (Z ₀ , Zi, Z ₂ , Z ₃ , z ₄ ) at a receiver 120 for more than 2 ^B different signal combinations, the N 'parallel signals (uo, Ui , U ₂ , U ₃ , U ₄ ) at an output of the signal encoder (110) to obtain a transmission result. The method 1000 further comprises, in a second step 1020, setting the mapping rule based on the transmission result.

Fig. 10B shows a flowchart of a method of configuring a signal encoder for generating a plurality of single modulation symbols based on B information bits. It is assumed here that the signal coder has, for example, the structure described with reference to FIGS. 6B or 7B. Thus, it is assumed that the signal encoder comprises vector images 680, 780 for non-linear mapping of possible bit patterns of the B information bits (ci, C2, ..., C _B ») and (ci, C ₂ ,..., C _B ) comprises 2 ^B different combinations consisting of at least two single modulation symbols (xi, X ₂ , ... _/ X _N ) or (xi, X ₂ , ..., XN ') according to a mapping rule, wherein the 2 ^B different combinations represent a true subset of a total modulation alpha M. The total modulation alphabet M is a cross product of sets M ₁ , MΣ ...., M _N > of actually occurring modulation values of the at least two individual modulation symbols (xi, x ₂ , X ₃ ..., X _N '). educated-. The method according to FIG. 10B is designated 1050 in its entirety. In a first step 1060, the method 1050 comprises determining received signals (yi, y ₂ ,..., Y _M ) or (yi, y ₂ ,..., Y _M ') or of quantized received signals ( zi, Z ₂ , ..., z _M ) or (zi, Z ₂ , ..., z _w ) for more than 2 ^B different combinations of the individual modulation signals {xi, x ₂ , ...., x _N ) or (xi, x ₂ , ..., X _N -) at an output of the signal encoder to obtain a transmission result. The method 1050 further comprises, in a second step 1070, setting the mapping rule based on the transmission result.

In the following, a further algorithm for determining the (redundant) room code will be described with reference to FIGS. 10c and 10d.

For example, said algorithm may be used to obtain M and a new modulation alphabet M 'from a total modulation alphabet. In other words, the method described below for determining a new modulation alphabet M 'can be used, for example, instead of the method described above for determining a new modulation alphabet M'. It is assumed here that Mi, M ₂ , ..., M _N > W describe single modulation alphabets.

The method according to the invention is designed to be used both in connection with a line-based communication system, for example according to FIGS. 1 and 2, and in conjunction with a wireless or wired communication system according to FIGS. 6a, 6b, 7a, 7b, 8. The single modulation alphabets M ₁ , M ₂ , ..., M _N / can for example be identical to the sub-modulation alphabets or single-modulation alphabets, as described with reference to FIGS. 6a, 6b, 7a and 7b. In other words, the single modulation alphabets M ₁ , M _Σ , ...., Mw may, for example, comprise quantities of possible single modulation symbols that can be transmitted over a channel of the communication system. In summary, M ₁ , M _Σ , ..., M _N 'are the N' single modulation alphabets.

Further, Z ₁ , Z _Σ ,.., Z _M 'M' are single quantization alphabets. In other words, Z ₁ , Z ₂ , ..., Z _M > describe the possible values at the outputs of the threshold value decoder or quantizer (see threshold value decider or quantizer 150 according to FIG. Transducer 640, 790 according to Figures 6a, 6b, 7a, 7b).

Thus, for output values Z ₀ , Zi, z ₂ , Z ₃ , z ₄ and Zi, Z ₂ ,..., Z _M and Zi, Z ₂ ,..., Z _M ', respectively:

Zi e Z ₁ , Z ₂ e Z ₂ , ..., -Lw e Z _M ..

That is, at least one of the single quantization alphabets (but preferably each of the single quantization alphabets) describes the possible values at the output of the respective quantizer 150_0, 150_l, 150_2, 150_3, 150_4; 440_l, 640_2, ..., 640_N; 790_l, 790_2, ..., 790_N.

Further, let Z = Z \ x Z ₂ x • • • • x% w be a total quantization alphabet. In other words, it holds:

z = [z _λ , z ₂ , .... z _M ] <E Z.

In the following, a method according to the invention for determining the redundant room code will be described step by step. 1. Set M = M ₁ x M ₂ x ... x M _N >, where | M | > 2 ^B.

The method according to the invention or the algorithm according to the invention for determining the (redundant) room code thus comprises, in a first step, determining one total modulation alphabet M. For this purpose, the following is set:

M = N ₁ x M ₂ x .... x M _N '.

In other words, an overall modulation alphabet M is determined to be a cross product of the N 'single-modulation alphabets.

For a width \ M \ of the total modulation alphabet M,

| * 4> 2 ^B.

2. For all z e Z: determine the i (z) for which:

Λ / 3x (z) = arg mxeaΛ / x Pr [z | x]

In other words, it is determined from the set M ₁ from the total modulation alphabet, that element i for which a conditional probability Pr | z | xJ that based on the combination x of transmitted single modulation signals (or based on a transmitted combination of signals) the combination z of received quantized modulation symbols arises, is maximized.

For this determination, it is preferred for a plurality of transmitted combinations x of single modulation symbols (or of, for example, binary modulation symbols)

Send signals) the conditional probability Prjzjx] for an occurrence of the received quantized combination z of modulation symbols. Then, in a preferred embodiment, the transmitted modulation symbol x is determined as the estimated value i (z) (associated with the received combination of quantized single modulation symbols z) for which said conditional probability with respect to the remaining possible modulation symbols of the overall modulation alphabet M assumes an above-average, but preferably a maximum value.

In other words, an estimated value i (z), which corresponds to an estimate of a good or optimal estimator that receives the combination of quantized received single modulation symbols z, is thus determined. The estimated value i (z) thus describes, at least approximately, (or provides an estimate) which combination of single modulation symbols was sent with a highest probability if the receiver z is the combination of received, quantized modulation symbols.

In other words, according to an estimation rule i (z), at least a plurality of combinations z of received quantized single modulation symbols (or combinations of received quantized signal combinations), but preferably each combination z of received quantized combinations of single modulation symbols (from the Ge - velvet quantization alphabet Z) is assigned an estimated value i for a transmitted combination of individual modulation symbols on which the received quantized combination of individual modulation symbols z is based.

3. For all xe M determine the following quantity:

In other words, for a plurality of combinations x (preferably for all combinations x out of M) of transmitted single modulation symbols, associated quantities S (x) associated with an associated transmitted combination x of single modulation symbols and all include those received quantized combinations of single modulation symbols z from the total quantization alphabet Z, which are mapped to a different than the transmitted combination x according to the estimation rule.

4. Determine x 'e M, for which:

Af 3 X ^f = arg mxeaΛ / x Y ^* Pr [z | x] zeS (x)

In other words, a transmitted combination x 'of single modulation symbols (from the base set M) identified with an above-average, preferably maximum, probability (relative to the other transmitted combinations x of single modulation symbols from the set M) is identified a resulting quantized combination z of single modulation symbols mapped to a false estimate i (z) according to the estimation rule (described in 2. above) (where the set S (x) is the incorrectly associated combinations associated with the combination x of received and quantized single modulation symbols). It is also assumed that an estimated value MzfX _gtsent e) is false if: * ( ^z ( ^x _{g «ends} )) ^{≠ x} _{ge« 7! Det} • ^In other words, an estimate is incorrect when the _e from the received combination "mpf _at g _s of quantized individual modulation symbols estimated combination x \ ^ _{mp / imgeπ)} does not match the actually transmitted combination ^ _sent ^from single modulation symbols on the received combination" received _at gene is consistent.

For this purpose, a sum of the individual probabilities is formed, each of which describes the probability with which a considered combination x ± of transmitted single modulation symbols mapped to received quantized combinations ze S (x), which are described in accordance with Regulation for determining an estimated value i (z (x _r )) are mapped to an estimated value i, so that i ≠ x ,.

In other words, from the overall modulation alphabet M considered (or from a reduced modulation alphabet M derived therefrom), that combination x 'of individual modulation symbols is identified that is suitable for deriving a correct estimated value x (z ( τ!)) brings a maximum unreliability (or at least an above-average unreliability) with it.

In other words, from the set M, the "worst" element x 'is identified, ie the transmitted combination x' of single modulation symbols, which in a receive-side estimation based on a received quantized combination z of single modulation symbols has a minimum or at least below average reliability.

In other words, the transmitted combination x ^r of single modulation symbols is identified on the receiver side with a lowest (or at least a below-average) reliability, based on the elements of the set M, for example, based on the resulting resulting received quantized combination of single modulation symbols, can be correctly reconstructed.

5. M <- M \ {x ¹ }, ie remove x 'from M.

In other words, the identified "worst" combination x 'of single modulation symbols is removed from the overall modulation symbol M to obtain a new, reduced modulation alphabet M.

Incidentally, the new scaled-down modulation alphabet M is used in a possible repetition of steps 2 to 5 instead of the overall modulation alphabet. Thus, the modulation alphabet M is reduced more and more as you repeat steps 2 through 5.

6. As long as \ M \> 2, continue at 2.

In other words, the method starts from step 2 ("For all z e Z: determine the x (z) for which:

... "), where the repetition uses the modulation alphabet reduced by the unreliable combination x 'of single modulation symbols identified in the fifth step instead of the original total modulation alphabet M.

In this way, the modulation alphabet M is recursively reduced in size until, for example, the modulation alphabet M has a width of 2 ^B or less than 2 ^B. For example, the procedure is repeated recursively as often as required until the scaled down Quantity M includes as many elements as are transmitted simultaneously across the channel.

Thus, a scaled-down modulation alphabet M, which is based on the original overall modulation alphabet M = Mi x M _Σ x .... x M _{1J ',} is obtained.

The reduced modulation alphabet M _ve rki _e inert is then used to set the mapping rule of the vector images 140, 680, 780 according to the invention.

In other words, the vector images 140, 680, 780 are set, for example, to downsize the 2 ^B combinations of information bits onto the 2 ^B elements of the reduced modulation alphabet ^. Thus, the method of the invention describes a method for determining which of the more than 2 ^B possible combinations of single modulation symbols are used by the vector images 140, 680, 780.

Incidentally, the method according to the invention can be modified in various ways. For example, in the fourth step, a plurality of unreliable combinations of single modulation symbols Xi, Xz ₁ ... Xp may be determined. In this case, the plurality of particular unreliable combinations of single modulation symbols in the fifth step may be obtained in common from the overall modulation alphabet M or from a reduced modulation alphabet M ^f obtained in a previous pass of the algorithm.

Further, other termination conditions may be used for the method. For example, in the fourth

Step calculated probability ^ Pr [z | x] as a

K = S (X)

Abort criterion be used. For example For all xe Af said probability less than or equal to a predetermined probability threshold, the method can be aborted, and it can be assumed that all contained in the reduced modulation alphabet M combinations of single modulation symbols are sufficiently reliable or can be detected at the receiver side with sufficiently high reliability.

10c furthermore shows a flow diagram of a method according to the invention for configuring a signal coder 110 for generating N 'parallel binary signals (Co, Ci, C ₂ , C ₃ , C ₄ , Uo, Ui, u ₂ , U ₃ , U ₄ ) based on B information bits (bo, bi, b2, b _β ). It is assumed that the signal encoder comprises a vector image for non-linear mapping of possible bit patterns of the B information bits to 2 ^B signal combinations on the N 'parallel binary signal, where N' is greater than B. The transmitted signal combinations are typically transmitted via a channel to a receiver, so that the receiver receives received, quantized receive signal combinations.

The method 1080 according to FIG. 10c comprises in a first step 1082 a determination of an unreliable transmission signal combination of the N 'parallel signals from a set of possible transmission signal combinations which have an above-average probability (or a maximum probability) ( for example, based on the other transmission signal combinations from the set M) results in a wrong estimate for the signal combination. In other words, a transmission signal combination x 'is determined which (for example in the presence of statistical signal interference during transmission) has a certain (typically comparatively high) probability of an incorrect estimated value i generated at a receiver, leads. In a second step 1084, the transmission signal combination x determined in the first step 1082 is removed from the set M of possible transmission signal combinations in order to obtain a smaller quantity M of possible transmission signal combinations.

Thereafter, the first step 1082 and the second step 1084 are repeated one or more times using the reduced amount M of possible transmit_signal combinations obtained in the second step 1084 of removing as the set of possible transmit-signal combinations used in the determination. In other words, the amount M is gradually decreased in repeating the steps 1082, 1084.

The method 1080 is therefore preferably a recursive method. A termination of the method 1080 takes place, for example, if the reduced quantity M is sufficiently small, for example has the width 2 ^B or a lower thickness.

10d further shows a flow diagram of a method according to the invention for configuring a signal coder for generating a plurality of individual modulation symbols Xi, X ₂ ,... X _N; Xi, X ₂ f ••• X _N ') based on B information bits (ci, C2 _' • -, c _B ', ci, Cz, ... C _B ). It is assumed that the signal encoder comprises a vector imager for non-linear mapping of possible bit patterns of the B information bits to 2 ^B different transmit combinations consisting of at least two single modulation symbols according to a mapping rule. It is assumed that the 2 ^B different combinations represent a true subset of a total modulation alphabet, where the total modulation alphabet is represented by a cross product of sets (eg, M w, M 1,... M _N ) of actually occurring ones Modulation values of the at least two individual modulation symbols is formed. It is further assumed that the signal encoder is designed is to transmit over a channel, the combinations of single modulation symbols to a receiver, so that received from the receiver, quantized receive combinations of single modulation symbols arise.

The method 1090 according to FIG. 10d comprises, in a first step 1092, determining at least one unreliable transmission combination x 'of individual modulation symbols from a set M of possible transmission combinations of single modulation symbols having an above-average probability (or a maximum probability in the With respect to other possible transmission combinations of single modulation symbols from the set of possible transmission combinations) in a false estimate i for a combination of transmitted single modulation symbols. An estimated value i indicates, for example, a transmission combination which, among all transmission combinations, leads most likely to a specific reception combination.

The method 1090 further comprises, in a second step 1094, removing the transmit combination x 'determined in the first step 1092 from single modulation symbols of the set M of possible transmit combinations of single modulation symbols by a reduced set M of possible transmit combinations of single modulation symbols to obtain.

The method 1090 further comprises repeating steps 1092, 1094 of determining and removing once or more than once. Here, the reduced quantity M of possible transmission combinations of single modulation symbols obtained in the step 1094 of removing is used in repeating the first step 1092 of determining as a new set of possible transmission combinations of single modulation symbols. In other words, within the scope of the method 1090, the amount M is reduced stepwise. Initially, the set M uses the total modulation alphabet, from which at least one (possibly also several) unreliable transmit combination x 'of single modulation symbols is then removed in step 1094 in order to obtain a reduced set M. In a subsequent re-execution of step 1092, the reduced amount M is used.

It should be noted, moreover, that the methods according to the invention can be used both for setting the imaging specification of the vector images used in the signal encoders according to the invention and for adjusting the mapping rule of the vector images used in the signal decoder according to the invention.

It should be noted, by the way, that x (^) denotes, for example, a transmission combination which, with all possible possible combinations of sans, results with the highest probability in the received quantized combination ^z of single modulation symbols. The set S (x) describes, for example, a set of received, quantized combinations of individual modulation symbols, which are mapped by an estimate, that is to say by forming an estimated value i (z), to an estimated value * which differs from the value of the transmission combination x different.

It should also be noted that in place of the combinations of single modulation symbols, in each case (for example, binary) signal combinations can occur if, for example, a wired transmission system according to FIGS. 1 or 2 is used.

7 also shows a flow diagram of a method according to the invention for configuring a signal decoder to generate B information bits from N 'parallel binary signals. The method can be used, for example, in Connected to a signal decoder 120, as has been described with reference to FIGS. 1 to 5. Thus, it is assumed that the signal decoder 120 includes a vector mapper 152 for non-linear mapping of signal combinations of the N 'parallel binary signals (zo, Zi, τ ₂ , Z ₃ , z ₄ ) to combinations of B information bits

(b ₀ , 0 ^ b ₂₉ O ₃ ), where: N '> B. It is further assumed that the vector image 152 is designed such that different of the signal combinations of the N' parallel binary signals (zo, Zi, Z ₂ , Z ₃ , Z ₄ ) are mapped to equal combinations of the B information bits so that a total amount of data output from the vector image per unit time is less than one input per unit time in the vector images over the N 'parallel binary signals amount of data. The method according to FIG. 11 is denoted in its entirety by 1100. The method 1100 includes, in a first step 1110, sending 2 ^B different signal combinations to the signal decoder. The method further comprises, in a second step 1120, analyzing the receive signals (yo, yi, y2 / y _3f y < ₁ ) ₁ that result at the input of the signal decoder in order to obtain an analysis result. The method 1100 further comprises, in a third step 1130, setting the mapping rule depending on the result of the analysis.

FIG. IIB shows a flow diagram of a method according to the invention for configuring a signal decoder to generate B information bits based on a plurality of single modulation symbols. The method according to FIG. IIB can be used for example with a signal decoder 820 according to FIG. 8. It is hereby assumed that the signal decoder 820 comprises a vector mapper 890 for non-linear mapping of combinations consisting of at least two single modulation symbols to combinations of B bits (ci, C ₂ ,..., C _B ) according to a mapping rule. It will Furthermore, it is assumed that the vector mapper 890 is designed such that different ones of the combinations of the single modulation symbols (zi, Z ₂ , ..., z _N ) are mapped to equal combinations of the B information bits and that the signal decoder is designed to be internally Combinations of the single modulation symbols to be distinguished from a Gesamtmodulationsapha- bet, wherein the total modulation alphabet comprises more than 2 ^B different combinations of single modulation symbols (zi, Zz, ...., z _N ).

The process according to FIG. IIB is designated 1150 in its entirety. The method comprises, in its first step 1160, transmitting 2 ^B different combinations of single modulation symbols Xi, X ₂ , ...., Xu to the signal decoder. The method 1150 includes, in a second step 1170, analyzing the receive signals resulting at an input of the signal decoder 820 to obtain an analysis result. The method 1150 further comprises in a third step 1180 for setting the mapping rule depending on the analysis result.

It should be noted, moreover, that the methods 1000, 1150, 1100, 1150 can be supplemented or supplemented by all those steps and features which, in the context of the devices according to the invention, with regard to a setting of the mapping rule or with regard to a selection of the modulation alphabet.

The devices according to the invention and the methods according to the invention can be implemented in hardware or in software. The implementation may be on a digital storage medium, such as a floppy disk, a CD, a DVD, a ROM, a PROM, an EPROM, an EEPROM, or a FLASH memory with electronically readable control signals that may interact with a programmable computer system. that the corresponding procedure is carried out. In general, the present The invention thus also in a computer program product with program code stored on a machine-readable carrier for carrying out one or more of the inventive methods when the computer program product runs on a computer. In other words, the invention can thus be realized as a computer program with a program code for performing at least one of the methods when the computer program runs on a computer.

The present invention will be summarized briefly below. With the aid of the present invention, reliable digital communication via multi-channel systems with coarse received signal quantization (analog / digital conversion) is made possible. Multi-channel systems are common in practice, such as in a wireline communication system which uses multiple lines in a cable simultaneously, or in a radio transmission system in which multiple antennas are used simultaneously at both the transmitter and the receiver.

A communication via a multi-channel system with coarse analog-to-digital conversion at the receiver is not possible according to the current state of the art and is made possible by the invention described first. This is because relevant methods of digital communication technology depend on high-resolution analog-to-digital conversion at the receiver. However, the resolution of analog-to-digital (A / D) converters is limited. Above all, very high-speed A / D converters are only available with relatively low resolution. Especially with multi-channel systems, it may therefore happen that a high-resolution A / D converter for the required transmission rate is not available.

The invention described here makes it possible to dispense with high-resolution A / D converter. It is even possible to establish a successful communication with the largest possible resolution of only one bit. In the present description, in addition to a sketch of the technical environment, both the technical problem and its solution have been described by the invention.

In summary, it can thus be stated that the present invention provides essential system components (for example a signal coder and a signal decoder) which enable reliable communication via a wired or wireless channel, even in the presence of a receive-side coarse quantization. The present invention thus provides a solution to the technical problem of employing appropriate signal processing and channel coding techniques for successful communication, despite the lack of high-resolution A / D converters. In other words, the present invention solves the technical problem of making relevant methods of channel coding and signal processing, which conventionally require high-resolution A / D conversion, also usable with coarse A / D conversion.

bibliography

[1] J.G. Proakis, Digital Communications, 3rd Edition,

McGraw-Hill, ISBN 0-07-113814-5, 1995.

[2] I.E. Telatar, "Capacity of Multi-antenna Gaussian Channels," European Transactions on Telecommunications, 10 (6): 585-596, November 1999.

[3] M. Joham, W. Utschick, and J.A. Nossek. "Linear trans-processing in MIMO Communications Systems", IEEE Transactions on Signal Processing, 53 (8): 2,700 2,712, August 2005.

[4] R.K. Y. Jiang and A.C. Singer, "On the separability of demodulation and decoding for multiple-antenna block-fading channel communications," IEEE Transactions on Information Theory, 49 (10): 2,709-2,712, October 2003.

Claims

claims

A signal encoder (110) for generating N 'parallel binary signals (Co, Ci, C2, C ₃ , C ₄ , Uo, Ui, U ₂ , U ₃ ,

U ₄ ) based on B information bits (bo, bi, b ₂ , b ₃ ), having the following features:

a vector image (140) for non-linear mapping of possible bit patterns of the B information bits to 2 ^B signal combinations on the N 'parallel binary signals,

where N 'is greater B.

2. The signal encoder (110) according to claim 1, wherein the 2 ^B signal combinations of the 2 ^{N '} total possible signal combinations on the N' parallel signals (Co, Ci, Cz, C ₃ , C ₄ , Uo, Ui, U ₂ , u ₃ , U ₄ ) are selected so that the 2 ^B different signal combinations after a transmission of the 2 ^B different signal combinations via a channel (130), in which a (J- brechen between the parallel signals (130_0, 130_l, 130_2, 130_3, 130_4), leading to received signals from which the information bits (bo, bi, b ₂ , b ₃ ) can be reconstructed using threshold value decisions (150_0, 150_l, 150_2, 150_3, 150_4).

A signal encoder (110) according to claim 1 or 2, wherein the vector mapper (140) is arranged so that a signal value of at least one of the parallel binary signals (Co, Ci, C2, C ₃ , C ₄ ; u ₀ , Ui, U2, U _3, U ₄₎ (of at least two of the B information bits bo, bi, b _2, b ₃₎ depends.

The signal encoder (110) according to any one of claims 1 to 3, wherein B is greater than or equal to 2 and N '= B + 1.

5. Signal encoder (110) according to one of claims 1 to

4, where B is greater than or equal to 2 and B + 1 ≦ N '≦ 2 ^{8 "1} .

6. Signal encoder (110) according to one of claims 1 to

5, in which vector images (140) have a look-up table, according to which possible bit patterns of the B information bits (b ₀ , bi, b ₂ , b ₃ ) signal combinations on the N 'parallel binary signals (Co, Ci, C ₂ , C ₃ , C ₄ , uo, Ui, U ₂ , U ₃ , U4) are assigned.

A signal encoder (110) according to claim 6, wherein the look-up table is hardwired or is programmable just once.

8. Signal encoder (110) according to one of claims 1 to

6, wherein the vector imager (140) comprises a memory circuit (450), the memory circuit being adapted to receive the B information bits (bo, bi, b2, b ₃ ) in parallel as address information for selecting a memory field, and the in the information stored in the selected memory field as a signal combination on the N 'parallel binary signals (Co, Ci, C ₂ , C ₃ , C ₄ ).

9. Signal coder (110) according to one of claims 1 to 8, wherein B equals 4 and N '= 5, and wherein the 2 ⁴ signal combinations on the five parallel binary signals Co, Ci, C ₂ , C ₃ , C ₄ following signal combinations [Co _/ Ci, C ₂ / C _3, C4] include:

[00000], [00001], [00010], [00100], [01000], [10000], [00011], [00101], [01001], [10001], [00110], [01010], [01100 ], [10010], [00111], [OH10].

A signal coder (664; 764; 810) for generating a plurality of single modulation symbols (X ₁ , χ ₂ , ..., x _N ; X _I , X ₂ , ..-, XN ¹ ) based on B information signals. Bits (ci, C ₂ , ...., c _B ', C _I , C ₂ , ..., C ₈ ), having the following characteristics:

a vector mapper (680; 780; 840) for non-linear mapping of possible bit patterns of the B information bits to 2 ^B different combinations consisting of at least two single modulation symbols, the 2 ^B different combinations representing a true subset of a total modulating alphabet (M) , where the total modulation alphabet (M) is formed by a cross-product of sets (Mi, M ₂ ,..., Λf _N ) of actually occurring modulation values of the at least two individual modulation symbols.

11. signal encoder (664; 764; 810) according to claim 10, wherein the ^B 2 various combinations so from the overall modulation alphabet (M) are selected such that the ^B 2 various combinations of transmission by an over- (through a passage 630; 784, 850, 830, 870), in which an influence between the individual modulation symbols (Xi, X2,..., XtJ; Xl, X2,..., XN _' ) follows, to receive signals (yi, y ₂ , ..., y _M , ^* Zi, Z ₂ , ..., z _M ; yi, y ₂ , ..., y M-; z _x , z ₂ , ...., z _{M '} , - y _x , y ₂ , ♦ ••, yN, Zi, Z2, ♦ •., ZN), from which the information bits (Ci, c ₂ ,..., c _B ; C i, C ₂ ,. .., c _{B '} ) are reconstructable using a threshold decision (640; 790).

A signal encoder (664; 764; 810) according to claims 10 or 11, wherein the signal encoder is arranged to provide the combination of single modulation symbols (xi, x ₂ , ..., x _N ; X _I , x ₂ ,. ..., XN>) for a simultaneous control of a plurality of channels of a modulator (850), wherein the single modulation symbols comprise complex-valued modulation values.

A signal encoder (664; 764; 810) according to any one of claims 1 to 12, wherein the vector imager (680; 870; 840) comprises a look-up table indicating an assignment of the possible bit patterns of the B information bits (C ₁ , Cz, ..., c _{B '} ,' C ₁ , C ₂ , • -., c _B ) Combinations consisting of at least single modulation symbols (Xi, X2, ..., XN; Xl, X2 / ••• _< XN ' ), where a combination of V single-modulation symbols V comprises real-valued or complex-valued modulation values.

14. A signal encoder (664, 764; 810) according to any one of the claims 10 to 13, wherein the vector image (680;

780; 840) comprises a memory array, the memory array being arranged to parallel the B information bits (Ci, C2, ..., c _{B '} , Ci, C2, ..., c _B ) as address information for selecting a memory array and the information stored in the selected memory field describing at least two complex-valued single modulation symbols (xi, x2, xi, xn, xi, x2, x, x, n) in parallel as the combination of single Output modulation symbols.

A signal encoder (664; 764; 810) according to any one of claims 10 to 14, wherein the vector mapper (680, 780; 840) is adapted to provide a single modulation value, the 2 ^B different combinations of information bits (ci, C ₂ , ..., c _{B '} , Ci, C2, ..., C _B ) assumes a total of W different values, is dependent on X information bits, where:

W < ^2X .

A signal encoder (664; 764; 810) according to any one of claims 10 to 15, wherein vector images (680; 780; 840) is adapted to output a total of N 'single modulation symbols x _N * in parallel to each bit pattern of the B information bits (Ci, C2e • -., CB-, CI, Cz, ..., c _B ), wherein an n-th single modulation symbol for the 2 ^B possible bit patterns total W _n . assumes different values, and where:

17. Signal coder (664; 764; 810) according to one of claims 10 to 16, with the following further feature:

a vector image imager (860) adapted to provide information about a state of a channel (830) between a transmitter (850) representing the combinations of the single modulation symbols (X ₁ , x ₂ , .. .., X _N ) emits in modulated form as a transmission signal, and a receiver (870), which generates a reception signal (y _lf y ₂ , _... , Y _N ) based on the transmission signal transmitted by the transmitter Information bits (Ci, C ₂ , ...., c _{B '} ) to be reconstructed from the received signal to receive, and

to adjust, based on the information about the state of the channel, the vector imager (840) so that the vector image maps bit patterns of the B information bits only to those combinations of single modulation symbols that result in received signals after the transmission over the channel What can be reconstructed by the receiver with at least a predetermined minimum reliability information bits.

18. Signal coder (664; 764; 810) according to any one of the claims 10 to 17, with the following further feature: a vector imager adjuster (860) adapted to provide information about a state of a channel between a transmitter adapted to represent the combinations of the single modulation symbols (xi, X ₂ , ..., x _N ) in a modulated form to be transmitted as a transmission signal, and a receiver (870) adapted to generate a reception signal (yi, y ₂ , ..., y _N ) based on the transmission signal transmitted from the transmitter to extract the information bits (ci , C ₂ , ..., c _B 0 from the received signal, to receive, and to adjust the vector imager (840) based on the information about the state of the channel so that the vector image bit pattern of the B information bits only on such combinations of single modulation symbols mapping, the transmission according to the Ü over the channel to receive signals (y i, Y2, - • • _t yi ") lead, the at least according to a difference measure a predetermined minimum difference.

19. The signal encoder (664; 764; 810) of claim 18, wherein the vector imager adjuster (860) is further configured to set the vector imager (840) based on the information about the state of the channel in that the vector map maps bit patterns of the B information bits (ci, C ₂ , ...., c _B ') only to those combinations of the single modulation symbols (Xi, X ₂ ,..., x _N ) that follow the transmission to receive signals (yi, y _Σ , ..., yN), so that at any two different bit patterns of the B information bits associated receive signals at least with respect to a received single modulation symbol (yi, yΣ, ..., y _N ) so that a probability that the received signals belonging to the different bit patterns of the B information bits for a given signal-to-noise ratio by a predetermined threshold decision rule to equal output Signals (Z ₁ , Z ₂ , ..., z _M ) of the threshold decision (880) are mapped, is smaller than a predetermined maximum probability.

A signal encoder (664; 764; 810) as claimed in any one of claims 10 to 19, wherein the vector mapper (680; 780; 840) is adapted to provide a combination of single modulation symbols (xi, x ₂ ,. .., X _N ) corresponding to output single modulation symbols in parallel.

The signal coder (664; 764; 810) according to any one of claims 10 to 20, wherein vector imagers (680; 780; 840) are adapted to provide bit patterns of the B information bits (ci, C2, ..., CB-; CI, C2, • -., C _B ) receive in parallel.

A signal coder (664; 764; 810) according to any of claims 10 to 21, wherein the vector mapper (680; 780; 840) is adapted to ensure that upon occurrence of an arbitrary sequence of multiple bit patterns, the B information bits (ci, C2, ..., c _{B '} , C i, C 2, ..., c _B ) only a total of 2 ^B different combinations of single modulation symbols (X ₁ , X ₂ , ..., X _N , "X _I , X _{2 Λ ''} -IX _{N '} ) occur at an output of the vector imager.

A signal encoder (664; 764; 810) according to any one of claims 10 to 22, wherein the signal encoder comprises a modulator (850) adapted to provide a combination of single modulation symbols (X ₁ , X ₂ ,

..., x _N ) from an output of the vector imager (840) and, based thereon, to generate a modulated time signal (830) whose time course is within a predetermined time interval of all B information bits (C ₁ , C ₂ , ...., c _B -) is dependent, so that the information of the B information bits is included in the time signal at the same time.

24. A signal encoder (664, 764; 810) as claimed in any one of claims 10 to 23, wherein the signal encoder comprises a modulator (850) adapted to output, from the vector imager (840), K combinations of single modulation symbols (840). xi, X ₂ , ...., X _N ) based on a sequence of K values of B information bits (Ci, C ₂ , ...., c _{B '} ) and based thereon by generating a Sequence of modulation states to produce a sequence of K modulated signal paths of a time signal, so that in the K modulated signal curves for an arbitrary sequence of information bits at most 2 ^B different modulation states from a Gesamtmodulationsalpha- bet of at least 2 ^B + 1 different Modulationszu- occur, wherein the modulator is designed to obtain a modulation state by a simultaneous one-to-one conversion of the individual modulation symbols of a combination of single modulation symbols in a time signal s.

25. A method (1000) for configuring a signal coder (110) for generating N 'parallel binary signals (Co, Ci, C ₂ , C ₃ , C ₄ , Uo, Ui, U ₂ , U ₃ , U ₄ ) based on B information bits (b ₀ , bi, b ₂ , b ₃ ), comprising a vector mapper (140) for non-linear mapping of possible bit patterns of the B information bits to 2 ^B signal combinations on the N 'parallel binary signals according to a mapping rule, where N' bigger B, with the following steps:

Determining (1010) of received signals (V _1, y _2, y _3, y _4; zo _r zif Z _2, Z _3, Z ₄₎ to a receiver (120) for more 2 ^B different signal combinations of N 'parallel signals on a Output of the signal encoder (110) to obtain a transmission result; and

Setting (1020) the mapping rule based on the transmission result.

26. A method (1000) according to claim 25, wherein 2 ^{B of} the different signal combinations are associated with combinations of B transmitted information bits (b ₀ , bi, b ₂ , b3), and wherein the remaining signal combinations represent invalid signal combinations.

27. Method (1000) according to claim 25 or 26, wherein the setting (1020) of the mapping rule is such that the information bits (bo, bi, b ₂ , b ₃ ) are obtained from the received signals (yo, yi, y ₂ , y ₃ ) are reconstructable using a threshold decision (150).

28. The method (1000) according to claim 27, further comprising the step of:

Applying a Threshold Decision to the Receive Signals (y ₀ , Ylr Y2r Y3r Y4) r UItI aUS to receive the receive signals threshold-valued receive signals (zo, Zi, Z2, Z3, Z4) corresponding to the more than 2 ^B different signal combinations of the N parallel signals (c ₀ , Ci, C ₂ , c _3f C ₄ ; u ₀ , U ₁ , u ₂ , U ₃ , U ₄ ) are assigned; and

wherein the setting (1020) of the mapping rule is such that the possible bit patterns of the B information bits (bo, bi, b ₂ , b ₃ ) such signal combinations on the N 'parallel binary signals (co, Ci, c ₂ , c _{3 /} C ₄ , Uo, Ui, U2, U ₃ , U ₄ ) are assigned from the more than 2 ^B different signal combinations on the N 'parallel signals, the distinguishable combinations of the threshold-valued received signals (zo, Zi, z ₂ ,. Z ₃ , z ₄ ).

The method (1000) according to any one of claims 25 to 28, wherein adjusting (1020) the mapping rule based on the transmission result is such takes place by mapping a set of 2 ^B signal combinations onto the possible bit patterns of the B information bits (b ₀ , b ₁ , b ₂ , b ₃ ) according to the mapping rule, by omitting signal combinations which become received signals (y ₀ , yi , y2r Yir Vi), which are unacceptably difficult to distinguish from received signals belonging to another signal combination in accordance with a predetermined distinguishing criterion, are obtained from a set of more than 2 ^B different total possible signal combinations.

30. Method (1000) according to claim 29, in which the distinguishing criterion is a measure of a difference between two different signal combinations of the N 'parallel signals (C ₀ , C ₁ , C ₂ , C ₃ , C ₄ , Uo, Ui, U2, U3, U4), two signals being considered to be unduly discriminable if the measure of the difference indicates that the difference is less than a predetermined minimum allowable difference.

31. Method (1000) according to one of claims 25 to 30, wherein the setting (1020) of the mapping instruction comprises the following steps:

Assigning an estimated transmit signal combination i (z) to a combination of (JE) of N 'received signals for a signal combination (ACI) of N' parallel signals (Co, C _x, C _2, C _3, C _4; Uo, Ui, U ₂ , U3, U ₄ ) at the output of the signal encoder (110);

Assigning a probability value ((J) (X ₁ )) to a signal combination (Xi) of the N 'parallel signals at the output of the signal encoder, wherein the

Probability value (φ (x ±)) is a statement about how high a probability that the estimated transmission signal combination i (JE) with the signal combination (x ±) matches the N 'parallel signals at the output of the signal encoder (1010), the probability value taking into account statistical uncertainties in generating the combination of the received signals from the signal combination of the N' parallel signals at the output of the signal encoder;

Repeating the steps of assigning an estimated transmit signal combination and assigning the probability value to obtain probability values for the more than 2 ^B signal combinations (xi); and

Setting the mapping rule so that possible bit patterns of the B information bits (B ₀ , Bi, B ₂ , B ₃ ) are mapped to signal combinations (Xi) whose associated probability values (φ (Xi)) are greater than a predetermined minimum probability value,

wherein the minimum likelihood value is greater than or equal to the smallest of a signal combination (Xi) of the N 'parallel signals at the probability value associated with the output of the signal coder.

32. A method (1050) of configuring a signal encoder (664; 764; 810) to generate a plurality of single modulation symbols (xi, xΣr> ..., XN; Xir * 2r ... x _N ») on B information bits (ci, c ₂ ,

..., c _B >; C _I , C ₂ , ...., c _B ), comprising a vector imager (680; 780; 840) for non-linear mapping of possible bit patterns of the B information bits to 2 ^B different combinations consisting of at least two single modulation symbols , according to a mapping rule, wherein the 2 ^B different combinations represent a true subset of a total modulation alphabet (M), where the is formed by a cross product of sets (Mi, M ₂ ,..., M _N ) of actually occurring modulation values of the at least two individual modulation symbols, comprising the following steps:

Determining (1060) received signals (yi, y _Σ , ..., YM, YI, y ₂ , ..., YM, ZI, Z ₂ , ...., z _M , Z i, Z ₂ , ... , z _M >) at a receiver for more than 2 ^B different combinations of the single modulation symbols at an output of the signal encoder to obtain a transmission result; and

Setting (1070) the mapping rule based on the transmission result.

33. The method (1050) according to claim 32, wherein ^B 2 of different combinations of the individual modulation symbols (xi, X2, .... XN; X _I, X2 _'' -, X _N ') combinations of the transmitted information bits B (C ₁ , C 2, ..., C _B , C _I , C 2, ..., c _{B '} ), and wherein the remaining combinations of the single modulation symbols represent invalid combinations.

34. The method (1050) according to one of claims 32 or 33, further comprising the following step:

Applying a threshold decision (640, 880) to the received signals (yi, y2, .-., YM / Yi / Y2r - • -, YM '/ Yif Y2r • _' •, YN) / by threshold-weighted received signals (Zi , Z2, •••, ZM / ZI, Z2, •••, z _{M '} , z \, zΣ, ..., z _N ) to obtain the more than 2 ^B different combinations, consisting of at least two Individual modulation symbols are assigned; and

wherein the setting (1070) of the mapping rule is such that the possible bit patterns the B information bits are associated with such combinations of the single modulation symbols that result in distinguishable threshold-valued receive signals.

35. The method (1050) according to one of claims 32 to 34, wherein the setting (1070) of the mapping rule is performed based on the transmission result such that the set of 2 ^B different combinations of single modulation symbols (X ₁ , x ₂ .

...., x _N ; X ₁ , X ₂ , ...., X _N ') to which the possible bit patterns of the B information bits (C ₁ , C ₂ , ..., C _B , C ₁ , C ₂ , ..., C _B' ) are mapped according to the mapping rule, by omitting combinations of the received signals (y _α , y ₂ , ..., y _M ; V ₁ , y ₂ , ..., y _M "; yif yzr - _" Yn ), which are unacceptably difficult to distinguish from received signals belonging to another signal combination according to a predetermined distinguishing criterion, is obtained from the set of more than 2 ^B total possible different combinations.

36. The method (1050) according to one of claims 32 to 35, wherein the setting (1070) of the mapping instruction comprises the following steps:

Associating an estimated combination of single modulation _symbols (i (z)) with a combination (%) of received signals (y _lf y ₂ , ..., VM; yi, y ₂ , •> -, y _M »; yi, y ₂ , ..., y ^) for a combination of single modulation symbols (Xi) at an output of the signal encoder (664; 764; 810).

Assigning a probability value (φ (Xi)) to a combination of single modulation symbols (xi) at the output of the signal encoder, wherein the probability value (φ (xi)) is indicative of how high a probability is in that the estimated combination of individual modulation symbols (i (z)) coincides with a combination of the single modulation symbols at the output of the signal encoder, the probability value providing statistical uncertainties in generating the combination of the received signals from the combination of the single modulation symbols considered at the output of the signal encoder;

Repeating the steps of assigning an estimated transmission combination and assigning a probability value to obtain probability values for the more than 2 ^B combinations of single modulation symbols; and

Setting the mapping rule such that possible bit patterns of the B information bits (ci, cz, ..., C _{B '} , Ci, c-2, ... _/ c _B ) are mapped to combinations (Xi) of single modulation symbols whose associated probability values (φ (Xi)) are greater than a predetermined minimum probability value,

wherein the minimum likelihood value is greater than or equal to the smallest likelihood value associated with a combination of single modulation symbols (Xi) at the output of the signal encoder.

37. _{Signal decoder} (120) for generating B information _bits (b _Q , b _ιy b ₂ , b ₃ ) from N 'parallel signals (y _o , yi, y ₂ , yo * Z ₀ , Zi, z ₂ , z ₃ , Z ₄ ), with the following features:

a vector image (152) for non-linear mapping of signal combinations of the N 'parallel signals to combinations of the B information bits (6 ₀ , Z> _t , £ ₂ , £> ₃ ) with N'> B, such that different ones of the signal combinations of the N 'parallel signals are mapped to the same combinations of the B information bits,

such that a total amount of data output by the vector imager per unit time is smaller than a data amount input per unit time in the vector imager via the N 'parallel signals.

38. The signal decoder (120) of claim 37, wherein the vector imager is configured to receive N 'parallel binary signals (zo, Z ₁ , z ₂ , Z ₃ , Z ₄ ) and signal combinations of the N' parallel binary signals, on a noise-free transmission of different combinations of B transmitted information bits (bo, bi, b _2, b ₃₎ 'based on the N lines, different combinations of the B empfan- genes information bits (b _o, b _v b _2, b ₃₎ to assign, and

around a considered signal combination of the N 'parallel binary signals, which is not based on a noise-free transmission of the different combinations of B transmitted information bits over the N' lines, that combination of B received information bits (^ _0> ^ i »^ _2> ^ ₃ ), which was transmitted in the presence of the considered signal combination with the highest probability over the N 'lines.

39. A signal decoder (120) according to claim 37 or claim 38, wherein the signal decoder comprises a threshold value discriminator (150) for selecting from the N 'parallel signals N' binary-valued threshold valued signals for use by the vector images (152) produce.

40. A signal decoder (120) according to claim 38 or 39, wherein the vector mapper (152) comprises a look-up table adapted to generate signal combinations of the N 'binary-valued threshold valued signals (z ₀ , Zi, Z ₂ , z _3f Z ₄ ) on combinations of the B received information bits (O ₀ , b _x , b ₂ , 0 ₃ ).

41. A signal decoder (120) according to claim 40, wherein the look-up table is hard-wired or is programmable just once.

42. A signal decoder according to claim 38, wherein the vector image (152) comprises a memory circuit (400) which is configured to generate the N 'parallel binary signals (z ₀ , zi, Z ₂ , Z ₃ , z ₄ ) as address information for selecting a memory array in parallel, and to output information stored in the selected memory array as the received information bits (ό _o , ό, ό ₂ , 6 ₃ ).

43. A signal decoder (120) according to any one of claims 1 to 42, wherein N '= 5, B = 4, and wherein the vector imager (152) is adapted to combine signals on the five parallel signals (z ₄ , Z ₃ , z ₂ , Zi, zo) according to the following mapping rule on four received information bits (6 ₃ , 6 ₂ , 0,, O ₀ ):

[0000]

[00001]

[0009]

[00100] [0011]

[0100]

[10000] [0101]

[0009]

[00111] -> [Olli]

[01001] -> [1000]

[10001] -> [1001] [00110] -> [1010]

[OHIO] -> [1011]

[01100] -> [1100]

[10010] -> [1101]

[01111] -> [HlO]

[Hill] -> [1111]

[0001]

[01010] -> [0010]

[01011] -> [0110]

[01101] -> [1000]

[10011] -> [0110]

[10100] -> [0011]

[10101] -> [1001]

[10110] -> [1010]

[10111] -> [Olli]

[11000] -> [0100]

[11001] -> [1000]

[11010] -> [1101]

[11011] -> [HH]

[11100] -> [0011]

[11101] -> [HH]

[11110] -> [HELL]

44. A signal decoder (820) for generating B information bits (c ,, c ₂ , ..., c _B. ) Based on a plurality of

Single modulation symbols (yi, y ₂ , ..., V _N ), with the following features:

a vector mapper (890) for nonlinear mapping of combinations (yi, y2, ···, ytj / to Z2f ··, z _N ) consisting of at least two single modulation symbols on combinations of B bits such that different ones of the combinations of Single modulation symbols can be mapped to the same combinations of B information bits.

45. The signal decoder (820) of claim 44, wherein the signal decoder is configured to internally combine The total modulation alphabet comprises more than 2 ^B different discrete combinations of single modulation symbols and to supply to the vector image (890) the internally distinguished combinations of single modulation symbols.

46. The signal decoder (820) of claim 44 or claim 45, wherein the vector imager (890) is arranged, and combinations of single modulation symbols (Yif Yii Y3r Yn; ZI, Z ₂ , Z ₃ , z _N ) based on a noise-free one Transmission of B transmitted information bits (Ci, C2, ...., C _B ') based, different combinations of the B received information bits (c ,, C ₂ , ..., c _B. ) Assign, and

a considered combination of single mod- ulation symbols (yi, y2, .. •, YN, Z \, "Z-2, • - • •, z _N ) that are not sent on a noise-free transmission of the different combinations of B. Information bits (ci, cΣfic _{B '} ) based, those

Combination of B received information bits

(C _p C ₂ , ..., c _B. ), Which in the presence of the considered signal combination ((yi, y2 / ••• / YN / ZI, Z2, • -., Z _N ) with a highest Probability was transferred.

47. signal decoder (820) according to any one of claims 44 to 46, wherein the signal decoder comprises a threshold decision-maker (880) in order from the received combinations (y _Σ y _lf, • • •, YN) of the individual modulation symbols combinations (Zi , Zz, ...., z _N ) of quantized or threshold-valued single modulation values.

48. A signal decoder (820) according to claim 47, wherein vector imager (890) is adapted to provide a combina- tion (z _\ , z ₂ , ..., Z _N ) of the quantized or threshold-valued single modulation values and received on a combination (c _p c ₂ , ..., c _B. ) of B received

Imaging information bits, wherein a value of at least one received information bits of more than a single modulation value (zi, Z ₂ , ...., z _N ) depends.

49. A signal decoder (820) according to one of claims 44 to 48, wherein the signal decoder is adapted to generate the single modulation symbols (yi, y ₂ , ..., y _N / z ₁ , Z ₂ , ..., Z _N ). from overlapping time periods or from a single time period of an input-time signal (830).

The signal decoder (820) of any one of claims 44 to 49, wherein the signal decoder (880) comprises a decoder configured to encode a n'-th single modulation symbol (yi, y ₂ , ..., yN) from W _n . assign possible values, where 1 ≤ n '≤ N, where N is a number of total processed single modulation symbols,

wherein the vector mapper (890) is adapted to receive the values associated with the single modulation symbols and to map them to combinations (c ,, c ₂ , ..., c _β .) of the B received information bits, where:

The signal decoder (820) of any of claims 44 to 50, wherein the signal decoder further comprises a vector image adjuster (894) configured to provide information about a state of a channel between a transmitter that is configured the combination (x _if x _2, ..., x _N) to emit the single modulation symbols in modulated form as the transmission signal (830), and the signal decoder (880) configured to generate a receive signal based on the transmit signal (830) transmitted by the transmitter to reconstruct, receive, and receive the information bits from the receive signal

to set the vector mapper (890) based on the information about the state of the channel such that the vector mapper (890) has combinations (yi, y ₂ , •••, YN / ZI, Z ₂ ,..., ZN ) of individual

Modulation symbols so on combinations (c ,, c ₂ , ..., c _B. ) Of received information bits maps that a combination of single modulation symbols (yi, y ₂ , ..., y _N ; Zi, Z ₂ ,. -., Z _N ), which is a transmitted combination of information bits assuming an undisturbed channel is assigned, is mapped to the transmitted combination of information bits, and that a combination (yi, y ₂ , ..., Y _N / zi , Z ₂ , ..., z _N ) of the single modulation symbols, which can not be assigned to a transmitted combination of information bits assuming an undisturbed channel, to a combination of information bits (c, c ₂ ,..., C _s ) is mapped to assuming a faulty channel with a maximum likelihood of the captured combination EMP -; can be assigned (yi, y _2, ••, YN zi, _Z2, ..., z _N) of individual modulation symbols.

52. A method (1100) of configuring a signal decoder (120) to generate B information bits

of N 'parallel signals (zo / Zi, Z ₂ , Z ₃ , Z ₄ ), with a vector mapper (152) for non-linear mapping of signal combinations of the N' parallel binary signals to combinations of the B information bits, with N '> B, see above that different ones of the signal combinations of the N 'signals are the same

Combinations of the B information bits so that one of the vector images per time The total amount of data output per unit of time in the vector imager via the N 'parallel binary signals is smaller than the amount of data input per unit time in the vector images, with the following steps:

Sending (1110) 2 ^B different signal combinations to the signal decoder;

Analyzing (1120) the received signals (yo, yi, Yit Y3r Y _A / Zor Zi, Z ₂ , Z3, Z4) at an input of the signal decoder to obtain an analysis result; and

Adjusting (1130) the mapping rule, depending on the analysis result.

53. The method (1110) of claim 52, wherein the 2 ^B different signal combinations are associated with combinations of B transmitted information bits (b ₀ , bi, b ₂ , b ₃ ).

54. The method (1100) according to claim 52 or 53, wherein the setting (1130) of the mapping rule is such that transmitted information bits (bo, bi, b ₂ , b3) from the received signals (yo, yi, Y2r Y3r y0 using a threshold decision in conjunction with a vector image are reconstructed.

55. The method (1100) according to one of claims 52 to 54, further comprising the step of:

Applying a threshold decision to the received signals (yo, Yi, Y2r Y3, YΛ) to obtain from the received signals 2 ^B combinations of threshold-valued received signals (zo, Zi, Z2, Z3, z ₄ ), the 2 ^B combinations of transmitted information bits (b ₀ , bi, b ₂ , b ₃ ) are assigned; and wherein the mapping rule is set (1130) such that the 2 ^B combinations of threshold-weighted received signals (zo, Zi, z ₂ , z ₃ , Zi) are mapped to associated combinations of information bits, and the remaining 2 ^{N '} - 2 ^B

Combinations of threshold-valued received signals to combinations of the received information bits (ό ₀ , ö ,, b ₂ , b ₃ ) are mapped, which are assigned in the signal space adjacent combinations of threshold-valued received signals based on an undisturbed transmission of the transmitted information bits (bo, bi, b ₂ , b ₃ ).

56. The method (1100) according to one of claims 52 to 55, wherein the setting (1130) of the mapping rule is such that signal combinations of the N 'parallel binary signals (Zo, Zi, Z2, Z ₃ , Z ₄ ), the noise-free transmission of different combinations of B transmitted information bits (bo, bi, b _Σ , b3) over the N 'lines, different combinations of the B received information bits (6 ₀ , ό ,, ά ₂ , ό ₃ ) are assigned, and

that a considered signal combination of the N 'parallel binary signals (z ₀ , Zi, Z ₂ , Z ₃ , Z ₄ ), which is not based on a noise-free transmission of the different combinations of B information bits (bo, bi, b ₂ , b ₃ ) over the N 'lines is based, which is assigned to each combination of B received information bits φ _o , b _iy b ₂ , b ₃ ), in the presence of the considered signal combination (z ₀ , Z ₁ , Z ₂ , Z ₃ , Z ₄ ) the highest probability has been transmitted over the N 'lines.

57. A method (1150) of configuring a signal decoder (820) to generate B received information tion bits (c ,, c ₂ , ..., c _B ,) based on a plurality of

Single modulation symbols (yi, Y2r .. •, Yu, zi, Z2, ..., z _N ), with a vector imager (890) for nonlinear mapping of combinations (yi, yΣ, - • -, Y ^ , 'Zi, Z ₂ , ..., z _N ) consisting of at least two individual modulation symbols on combinations of B received information _bits (c ,, c _2J ..., c _B. ) According to a mapping rule, so that different of the combinations of the single modulation symbols to equal combinations of the B received information bits

(c ,, c ₂ , ..., c _B. ), with the following steps:

Transmitting (1160) from 2 ^B different combinations (X ₁ , X ₂ , ..., x _N ; y i, y ₂ , ..., y u) of the single modulation symbols to the signal decoder;

Analyzing the receive signals (yi, y2,..., Yij) resulting at an input of the signal decoder to obtain an analysis result; and

Adjusting (1180) the mapping rule depending on the analysis result.

58. The method (1100) of claim 57, wherein the 2 ^B different combinations (Xi, x _Σ , ..., x _N ; Yi, Y _Ϊ , - •, Yn) of the single modulation _{symbols correspond} to different combinations of the B transmitted information _bits (ci, C _2f ..., c _N ) are assigned.

59. The method (1150) according to claim 57 or claim 58, wherein the setting (1180) of the mapping instruction is such that the transmitted information bits (ci, C ₂ , .. • _r c _B ) from the received signals (yi, yΣ / ..., y _N , Zi, z _{2 /} ..., z _N ) can be reconstructed using the mapping rule.

60. The method (1150) according to any one of claims 57 to 59, further comprising the step of:

Applying a threshold decision or a quantization to the received signals (yi, y ₂ ,..., Y _N ) in order to obtain from the received signals 2 ^B combinations (zi, Z2,..., Z _N ) of thresholded single modulation symbols associated with 2 ^B combinations (Ci, Cz, ..., c _{B '} ) of transmitted information bits; and

wherein adjusting (1180) the mapping rule is such that the 2 ^B combinations of threshold evaluated single modulation symbols (Z _1, Z _2, • -., z _N) (to corresponding combinations of received information bits c ,, c _2, ..., c _β .), and that remaining combinations of threshold-valued single modulation symbols (zi, Z 2,..., z _N ) that are not associated with a combination of transmitted information bits, are combined to a combination of received information _bits (c _ι , c ₂ , ..., c _B ) are mapped, which is associated with a signal space adjacent to the combination of threshold-valued single modulation symbols, which is based on a combination of transmitted information bits.

61. Method (1150) according to one of claims 57 to 60, in which the setting (1180) of the mapping rule is carried out such that combinations of the individual modulation symbols (yi, y2, ..-, yij; zi, Z2,. .., z _N ) based on a noise-free transmission of different combinations of B transmitted information bits (Ci, C ₂ , ..., c _B '), different combinations of B received information bits (c ,, c ₂ , ..., c _B. ), and that a considered combination (yi, yΣ, ..., yi *, Zi, Z ₂ , ..., Z _N ) of the single modulation symbols, not on one noise-free transmission of the different Combinations of the B transmitted information bits (ci, C ₂ , ..., c _{B '} ) based on which is associated with that combination of B-received information bits (c ,, c ₂ , ..., c _B. ) Present in the present one the considered combination of the single modulation symbols (yi, y ₂ , ..., y _N , Zi, Z ₂ , ..., z _N ) has been transmitted with a highest probability.

62. A method of operating a signal encoder (110) for generating N 'parallel binary signals (Co, Ci,

C2, _C3, C4; uo, Ui, U ₂ , U3, U ₄ ) based on B information bits (bo, bi _f b ₂ , b ₃ ), comprising the following steps:

Nonlinear, vectorial mapping of a possible bit pattern of the B information bits to a signal combination predetermined by 2 ^B on the N 'parallel binary signals, where N' is greater B.

63. A method of operating a signal encoder (664; 764; 810) to generate a plurality of individual

Modulation symbols (xi, X2, • • •, N; Xi, X2, • • •, _N X O, based on B information bits (Ci, C _2, ••, CB ', ^* CI, C2, •••. C _B ), with the following steps:

nonlinear, vectorial mapping of a possible bit pattern of the B information bits to an allowable combination given by 2 ^B consisting of at least two single modulation symbols (xi, x ₂ ,..., x _N ; X _I , X ₂ ,. X _N -), where the 2 ^B predetermined allowable combinations represent a true subset of a total modulation alphabet (M),

where the total modulation alphabet is represented by a cross product of sets {Mi / MΣ, ..., Mn '; Mi, MΣ, ..., M _N ) is formed by actually occurring modulation values of the at least two individual modulation symbols.

64. A method of operating a signal decoder (120) to generate B received information bits

(b _o , b _λ , b ₂ , b ₃ ) from N 'parallel binary signals (zo, Zi, Z ₂ , Z ₃ , Z ₄ ), comprising the following steps:

Nonlinear, vectorial mapping of a signal combination of the N 'parallel binary signals to a combination of B received information bits, with N'> B, according to a mapping rule, where, according to the mapping rule, different ones of the signal combinations of the N 'parallel binary signals onto equal combinations of B Information bits are mapped,

so that one per unit time by the vectorial

Mapping a total amount of data generated from the N 'parallel binary signals is less than a data amount delivered per unit time over the N' parallel binary signals.

65. A method of operating a signal decoder (820) to generate B received information bits

(C ₁ , C ₂ , ..., c _B. ) Based on a plurality of individual

Modulation symbols (yi, y ₂ , ... / YN, Zi, Z ₂ , - ♦ •, z _N ), with the following steps:

Nonlinear, vectorial mapping of a combination of at least two single modulation symbols (yi, yΣ, _'' , YN, ZI, Z2, •, ZN) to a combination of B received information bits (c ,, c ₂ , ..., c _B. ) according to a mapping rule,

wherein according to the mapping rule different combinations of the individual modulation symbols (y i, y _2, ..., YN, Z _1, Z _2, ..., z _N) on the same combinations of the B information bits (c ,, c _2,. .., c _B ,).

66. Computer program for carrying out a method according to one of claims 25 to 36, 52 to 65 or 71 to 81, when the computer program runs on a computer ter.

67. A data transmission system (100; 200) for transmitting B information bits (bo, bi, b ₂ , b ₃ ) from a data transmitter to a data receiver, having the following features:

a signal encoder (110) for generating N 'parallel binary signals (Co, Ci, C ₂ , C ₃ , Uo, Ui, U ₂ , U ₃ , U ₄ ) based on the B information bits, the signal encoder comprising a vector mapper (140 ) for non-linear mapping of possible bit patterns of the B information bits (bo, bi, b ₂ , b3) to 2 ^B signal combinations on the N 'parallel binary signals (Co, Ci, C2, C ₃ ; U ₀ , Ui, U ₂ , U3, U4), where N 'is greater B;

N 'transmission line (130_0, 130_l, 130_2, 130_3, 130_4) for transmitting the N' parallel binary signals (uo, Ui, U ₂ , U ₃ , U4) from a data transmitter to a data receiver; and

A signal decoder for receiving the N 'transmitted parallel signals (yo, yi, yΣ / y3, y4> from the N' transmission lines, and for reconstructing the B information bits from the received N 'parallel binary signals, to receive information bits

(b _Oi b _ιt b ₂ , b ₃ ), wherein the signal decoder

(120) a vector mapper (152) for nonlinearly mapping the signal combinations of the N 'received parallel signals (y ₀ , yi, Y2, Y3, Yi) or binary values derived therefrom by a threshold decision. ren signals (zo, z _\ z _Σ, zj, Z ₄₎ combinations of B received information bits (b _0, b _lt b _2, b _3),

wherein the vector image of the signal decoder is arranged so that different ones of the signal combinations of the N 'parallel signals (yo, yi, yΣ, Y3r Yn, Zo, Zi, Z ₂ , Z ₃ , Z ₄ ) are based on the same combinations of B

Information bits (O ₀ , 0 ₁ , 0 ₂ , 0 ₃ ), and that a total amount of data output from the vector image (152) per unit time is smaller than one per unit time in the vector images over the N 'parallel signals (y ₀ , yi, y2r y ₃ , y ₄ , z ₀ , Z ₁ , z ₂ , Z ₃ , Z ₄ ) entered amount of data.

68. The data transmission system according to claim 67, wherein a mapping rule of the vector image is obtainable by performing the following steps:

a) determining at least one unreliable transmit signal combination of the N 'parallel signals from a set of possible transmit signal combinations which results in an above-average probability in an incorrect estimate for the transmit signal combination, an estimate indicating a transmit signal combination among all transmit-signal combinations, most likely results in a particular receive signal combination (yo, yi, y2, y3, Yi);

b) removing the transmission signal combination determined in a) from the set of possible transmission signal combinations in order to obtain a reduced quantity of possible transmission signal combinations; and

c) Repeating the one or more times

Steps a) and b), wherein the reduced amount of possible transmission received in b) Signal combinations in a repetition is used as the amount of possible transmission signal combinations used in a),

until the reduced amount of possible transmission

Signal combinations a mapping rule of the vector image (140) determines.

69. A data transmission system for transmitting B information bits from a data transmitter to a data receiver, comprising:

a signal encoder (664; 764; 810) for generating a plurality of individual modulation symbols (xi, x ₂ ,..., x _N ; X _I , X ₂ , ..., Xw) based on the B information signals. Bits (ci, C ₂ , ...., c _{B '} , Ci, C ₂ , ..., c _B ), with a vector map (680; 780; 840) for non-linear mapping of possible bit patterns of the B information bits to 2 ^B different combinations consisting of at least two single modulation symbols, wherein the 2 ^B different combinations represent a true subset of a total modulating alphabet (M), where the total modulation alphabet (M) is represented by a cross product of sets (Mi, Af ₂ , .. ., M _N ) is formed by actually occurring modulation values of the at least two individual modulation symbols;

a channel for transmitting the plurality of single modulation symbols from a data transmitter to a data receiver; and

a signal decoder (820) for generating B information bits (C ₁₉ C ₁ c _B. ) based on the plurality of single modulation symbols (yi, y ₂ ,..., ytj) transmitted over the channel, with a vector image (890 ) for non-linear mapping of combinations (yi, y ₂ , ..., y _N , Zi, Z ₂ , ...., z _N ) consisting of at least two single modulation symbols to combinations of B bits such that different ones of the combinations of the single modulation symbols are mapped to equal combinations of the B information bits.

70. The data transmission system according to claim 69, wherein a mapping rule of the vector image of the signal coder is obtainable by performing the following steps:

a) determining at least one unreliable transmit combination of single modulation symbols from a set of possible transmit combinations of single modulation symbols resulting in an above-average probability in an incorrect estimate for the combination of single modulation symbols,

wherein an estimate indicates a transmit combination most likely to result in a particular receive combination Cy ₁ , Y ₂ , ••, yu among all transmit combinations;

b) removing the transmit combination of single modulation symbols determined in a) from the set of possible transmit combinations of single modulation symbols to obtain a reduced set of possible transmit combinations of single modulation symbols;

c) repeating steps a) and b) once or more than once, wherein the reduced quantity of possible transmission combinations of single modulation symbols obtained in b) in a repetition is the set of possible transmission combinations of individual modulation symbols used in a) is used, until the reduced amount of possible transmit signal combinations determines a mapping specification of the vector imager (140) of the signal encoder.

71. A method (1000) for configuring a signal coder (110) to generate N 'parallel binary signals (Co, Ci, C ₂ , C ₃ , C ₄ , u o, U i, U ₂ , U ₃ , U ₄ ) based on B Information bits (bo, bi, b ₂ , b3), comprising a vector image (140) for non-linear mapping of possible bit patterns of the B information bits to 2 ^B signal combinations on the N 'parallel binary signals according to a mapping rule, where N' is greater B,

wherein said signal encoder is designed to have a channel, the transmit signal combinations (c <j, c _x, C _2, C _3, C4; Uo, Ui, U _2, U 3, U ₄₎ to be transmitted to a receiver so that at the receiver received, quantized receive signal combinations (zo, Zi, Z2, Z3, Z ₄ ) arise,

with the following steps:

a) determining at least one unreliable transmission signal combination of the N 'parallel signals from one

Set of possible transmit-signal combinations that results in an incorrect estimate of the signal combination with an above-average probability

wherein an estimate indicates a transmission combination that is most likely to result in a particular reception combination among all transmission combinations;

b) removing the transmission signal combination determined in a) from the set of possible transmission Signal combinations to obtain a reduced set of possible transmit signal combinations; and

c) repeating steps a) and b) once or several times, wherein the reduced quantity of possible transmission signal combinations obtained in b) is used in a repetition as the set of possible transmission signal combinations used in a).

72. Method of Configuring a Signal Encoder

(664; 764; 810) for generating a plurality of single modulation symbols (X ₁ , X ₂ , ...., X _N ; X ₁ , X _2f ..., x _N ») based on B information bits (ci, C ₂ , ..., c _{B '} , C i, C ₂ , ...., c _B ), with a vector mapper (680; 780; 840) for non-linear mapping of possible bit patterns of the B information bits to 2 ^B various transmit combinations comprising at least two single modulation symbols, according to a mapping rule, wherein the 2 ^B different combinations represent a true subset of a total modulation alphabet (M), the overall modulation alphabet being represented by a cross product of sets (Mi, M Σ, • • •, M _N ) is formed by actually occurring modulation values of the at least two individual modulation symbols,

wherein the signal encoder is adapted to transmit over a channel the combinations of single modulation symbols to a receiver so that received, receive at the receiver, quantized receive combinations of single modulation symbols,

with the following steps:

a) determining at least one unreliable transmit combination of single modulation symbols from a Set of possible transmit combinations of single modulation symbols that results with an above-average probability in a wrong estimate for the combination of single-modulation symbols,

wherein an estimate indicates a transmission combination that is most likely to result in a particular reception combination among all transmission combinations;

b) removing the transmit combination of single modulation symbols determined in a) from the set of possible transmit combinations of single modulation symbols to obtain a reduced set of possible transmit combinations of single modulation symbols;

c) repeating steps a) and b) once or more than once, wherein the reduced quantity of possible transmission combinations of single modulation symbols obtained in b) in a repetition is the set of possible transmission combinations of individual modulation symbols used in a) is used.

73. The method of claim 72, wherein step a) comprises the steps of:

al) determining estimates (i (z)) describing estimated transmit combinations of single modulation symbols for a plurality of quantized receive combinations of single modulation symbols received at the signal decoder;

a2) determining quantities (S (x)) of received quantized combinations (x) of individual Modulation symbols for which an associated estimate (i (z)) describing an estimated transmitted transmission combination of single modulation symbols deviates from the actually transmitted combination (x) of single modulation symbols for a plurality of different actual transmitted combinations of Single modulation symbols (x); and

a3) determining a transmission combination (x ^r ) of transmitted single modulation symbols, such that at the transmitting end presence of the particular transmission combination (x ') of transmitted single modulation symbols a probability (^^ [zlx]) that the receiving side a quantized (x) sated combination of single modulation symbols from one of the quantities determined in a2) is above average.

74. The method of claim 73, wherein step al) comprises determining a combination of single modulation symbols as the estimated transmit combination (i (z)) of single modulation symbols such that for the estimated combination (x (z) ) of single modulation symbols, a conditional probability (P _r [Js | x]) for which an occurrence of a corresponding received quantized receive combination (JE) of single modulation symbols is larger than average.

75. The method according to claim 73 or 74, wherein the

Step a3) determining conditional probabilities (P _r [S | x]) for an occurrence of received quantized combinations (s) of individual modulation symbols from the set (S (x)) determined in step a2) below Requirement that a considered transmission combination (x) of sent Single Modulationssininbolen output by the signal encoder comprises,

and wherein step a3) further comprises summing conditional probabilities (P _c [z | x]) associated with the considered transmit combination (x) for a plurality of received quantized combinations (z) of single modulation symbols from that in step a2) comprises a certain amount (S (x)),

an overall probability of an amount (S (x)), determined for an occurrence of received quantized combinations (z) of individual modulation symbols, determined in step a2) for a viewed transmission combination (x) of transmitted individual - to get modulation symbols.

76. The method of claim 75, wherein step a3) comprises determining global probabilities for a plurality of considered transmit combinations (x) and selecting the transmit combination for which the overall probability for occurrence of received ones quantized combinations (z) of single modulation symbols from the determined in step a2) amount (S (x)} is above average.

77. The method according to one of claims 72 to 76, wherein the set of possible transmission combinations of single modulation symbols by repeating the

Steps a) and b) is recursively reduced until a reduced set of possible transmission combinations of single modulation symbols having a width of 2 ^{B is produced} .

78. The method according to any one of claims 72 to 77, wherein the set of possible transmission combinations of single modulation symbols by repeating the Steps a) and b) is recursively reduced until a probability that, when using a transmission combination of the recursively reduced set of possible transmission combinations, a probability for a receiver-side occurrence of a wrong estimate, the estimated transmission combinations of Describes individual modulation symbols, and which differs from the actually transmitted transmission combination, is less than or equal to a predetermined barrier.

79. The method of claim 72, wherein an initially used set of possible transmit combinations of single modulation symbols is a cross product of sets of possible single modulation symbols.

80. The method of claim 72, further comprising adjusting the mapping rule based on the recursively reduced set of single modulation symbols such that the vector imager selects bit patterns of the B information bits on transmit combinations of single modulation symbols from the recursive reduced quantity of single modulation symbols.

81. Method of Configuring a Signal Decoder

(120) for generating B information bits (b _o , b _λ , b ₂ , b ₃ ) from N 'parallel signals (z ₀ , Zi, Z ₂ , Z ₃ , Z ₄ ) with a vector image (152) for not linear mapping of signal combinations of the N 'parallel binary signals to combinations of the B information bits, with N'> B, such that different ones of the signal combinations of the N 'signals are mapped to equal combinations of the B information bits, such that one of the vector information bits Images per unit of time total amount of data is less than a data amount entered per unit time in the vector images over the N 'parallel binary signals,

wherein the signal coder is adapted to transmit the transmit signal combinations to a receiver over a channel such that quantized receive signal combinations received at the receiver are produced,

with the following steps:

a) determining at least one unreliable transmit-signal combination of the N 'parallel signals from a set of possible transmit-signal combinations that results in a false estimate of the signal combination with an above-average probability;

(b) removal of the transmitters specified in (a)

Signal combination of the set of possible transmit signal combinations to obtain a reduced set of possible transmit signal combinations; and

c) Repeating the one or more times

Steps a) and b), wherein the reduced amount of possible transmission signal combinations obtained in b) is used in a repetition as the amount of possible transmission signal combinations used in a).