WO2003084080A2

WO2003084080A2 - System and method for signal time alignment using programmable matched filter coefficients

Info

Publication number: WO2003084080A2
Application number: PCT/US2003/009235
Authority: WO
Inventors: Donald J. Rasmussen; Jeffrey C. Schmidt
Original assignee: Meshnetworks, Inc.
Priority date: 2002-03-27
Filing date: 2003-03-26
Publication date: 2003-10-09
Also published as: AU2003223354A1; WO2003084080A3; AU2003223354A8

Abstract

A method and system for time aligning a received transmitted signal with the receiver clock signal is disclosed herein. Time alignment is achieved by applying the proper matched filter coefficients (114) to account for the time difference between received transmitted signal and the receiver clock signal. For direct sequence signals, time alignment is achieved between the spreading chips. For non-direct sequence signals, time alignment is achieved between the symbols. Time alignment is obtained by applying the proper matched filter coefficients (114) to add or subtract propagation delay account for the time difference between the received transmitted signal and received clock signal. The shape of the received synchronization correlation curve is used to determine the selection of the proper matched filter coefficients (114) for fine timing correction. The matched filter coefficients are selected at synchronization and updated throughout the wave form reception. For direct sequence signals with rake receivers, timing alignment is easily applied to multiple rake taps (118).

Description

System and Method For Signal Time Alignment Using Programmable Matched Filter Coefficients

BACKGROUND OF THE INVENTION Field of the Invention:

[OOOl] The present invention relates generally to fixed or mobile digital communication systems, in particular, mobile wireless ad-hoc communication systems. More specifically, the present invention relates to a novel and improved system and method for obtaining and maintaining timing alignment between the received transmitted signal and the receiver reference signals of a digital communication system.

Description of the Related Art:

[0002] Many different types of wireless communication networks are in existence today.

Examples of these types of networks are terrestrial or satellite-based cellular telephone networks, terrestrial or satellite-based data communication networks, and radar networks, to name a few. These systems often employ spread spectrum modulation techniques to render the communications signals less susceptible to noise, interference, and multi-path channel effects.

[0003] Time aligning the received clock signal with the received transmit signal is critical to maximizing system performance for non-spread and spread spectrum systems. Proper time alignment increases the received signal-to-noise ratio by sampling the matched filter output at the optimum time. At the optimum sampling time, the matched filter maximizes the received signal power, while minimizing the received noise power. The optimum sampling time is achieved by aligning the received transmit signal with the receiver clock signal, which determines the sampling time of the received signal.

[0004] In the past, time alignment methods have focused on time aligning the receiver clock signal to the received signal using information generated from the received signal. Initial analog techniques used phase-lock loop (PLL) structures for adjusting the receiver clock signal. Digital techniques have also used a phase-lock loop structure for adjusting the receiver clock signal. Advances in digital signal processing capability have allowed more signal processing to be performed in the digital domain. For example, implementation of the matched filter has moved into the digital domain, especially for complicated pulse-shaping functions used to increase the transmitter bandwidth efficiency. Implementation of the time alignment method using these digital techniques have focused on control of the receiver clock signal, over-sampling techniques, and interpolation techniques. In all three of these approaches, the matched filter coefficients are not adjusted, but fixed.

[0005] The receiver clock signal correction approach (using a PLL) offers the advantage of operating at a lower clock frequency. Typically, the sample clock rate is lx or 2x the system symbol or chip rate. The matched filter operation is performed at this lower system clock rate. The additional hardware required to implement the phase-lock loop is the major disadvantage with this approach. This disadvantage is compounded when a RAKE receiver architecture is implemented, since each tap requires a phase-lock loop. [0006] The over-sampling approach provides timing alignment by selecting the appropriate sample from a set of over-sampled signals using a selection algorithm. A higher frequency sampling clock is used to over-sample the received signal and limits the time alignment accuracy. Typically, the sampling frequency is 2x to lOx the system symbol or chip rate. An advantage of this approach is that a phase-lock loop is not required. The major disadvantage is that the matched filter function is performed at a high clock rate, which increases complexity and power consumption. Since the over-sampling approach does not require a phase-lock loop, it is more easily applied to a RAKE receiver architecture. Time alignment for each RAKE tap is achieved by using a selection algorithm to select the proper sampling time for each rake tap from the over-sampled signal.

[0007] The interpolation approach provides timing alignment by using a poly-phase filter structure to reduce the timing alignment error from the matched filter. An advantage of this approach is that the matched filter can use a low clock rate, typically lx or 2x the system symbol or chip rate. Interpolating the matched filter output using a poly-phase filter structure reduces the timing alignment error. Furthermore, the poly-phase filter structure allows the interpolation operation to operate at the same clock rate as the matched filter. The disadvantage of the poly-phase filter structure is the hardware complexity required to perform each filter interpolation operation. However, by increasing the poly-phase filter clock rate and power consumption, hardware complexity can be reduced by performing multiple filter interpolation operations using the same filter structure with changing filter coefficients. Since the interpolation approach does not require a phase-lock loop, it is more easily applied to a RAKE receiver architecture. Time alignment for each RAKE tap is achieved by selecting the interpolation filter output for each tap from the poly-phase interpolation filter structure. However, a selection algorithm needs to be implemented for each tap, which can increase the complexity of the system.

[0008] Each of the time alignment techniques described above have unique advantages and disadvantages as stated above. In each of the above techniques the matched filter coefficients remain static in order to provide or support time alignment in the receiver. The matched filter performs a filtering function while another operation provides the time alignment. Also, each of the above techniques can be incorporated in a RAKE receiver architecture with the advantages and disadvantages discussed above.

[0009] Therefore, there is a need for an improved communication receiver capable of achieving optimum receive timing, while at the same time minimizing the necessary clock rate and reducing the hardware complexity. Furthermore, it would be advantageous for such a device to also work in a RAKE receiver architecture for multi-path channel mitigation in mobile and fixed communication environments.

SUMMARY OF THE INVENTION [OO10] It is therefore the object of the present invention to provide a time alignment approach that selects matched filter coefficients based on a timing selection algorithm to achieve optimum receiver timing, while minimizing clock rate and hardware complexity. Furthermore the principles of the present invention can be applied to a RAKE receiver architecture for multi-path channel mitigation in mobile and fixed communication environments.

[0011] The invention is embodied in a system and method for time alignment of a demodulator to a received signal using a selection algorithm based on the shape of the received correlation curve to select the matched filter coefficients used in correcting the timing error. The present invention also provides a novel and improved method and system for reducing the receiver clock rate and hardware complexity for time alignment for conventional and RAKE receiver architectures. The embodiments of the present invention are applicable to either parallel or serial demodulator architectures. The time alignment approach includes a matched filter section with programmable filter coefficients for timing control. A phase rotator is coupled to each of the matched filter outputs to enable parallel or serial demodulation. A SYNC and Serial Probe (SYNC/SP) processing section is coupled to the SYNC phase rotator.

[0012] The SYNC/SP processing section determines synchronization first during the SYNC section of the received waveform followed by the coarse time alignment. Coarse time alignment is set by the receiver clock rate based on the highest correlation peak levels during the SYNC and serial probe sections of the received waveform. Fine time alignment is achieved in the Correlation Function Comparison section by comparing the received correlation samples about the correlation peak with the expected correlation function. The coarse timing alignment sets the timing control on the demodulator with respect to the receiver clock. For a RAKE receiver architecture, the coarse timing alignment sets the timing for each of the RAKE tap processes in the demodulator. The fine timing alignment controls the matched filter coefficients used to provide the additional timing correction less than the receiver clock period.

BRIEF DESCRIPTION OF THE DRAWINGS [0013] These and other objects, advantages and novel features of the invention will be more readily appreciated from the following detailed description when read in conjunction with the accompanying drawings, in which:

[0014] Fig. 1 is a conceptual block diagram illustrating an example of a system for performing receiver clock time alignment by selecting the matched filter coefficients based on a selection algorithm, which uses a correlation curve;

[0015] Fig. 2 is an example of the sliding correlation function for a 1023 psuedo-noise (PN) code used by the system shown in Fig. 1 to establish timing alignment between the receiver signal and the receiver clock signal; [0016] Fig. 3 is an example of the sliding correlation function for the 1023 PN code with a direct and multi-path signal of equal amplitude at the receiver;

[0017] Fig. 4 is an example of the correlation and squared correlation curves associated with Minimum Shift Keyed (MSK) and Binary Shift Keyed (BPSK) modulated signals, where one of these curves are used to estimate the timing alignment depending on the receiver architecture used;

[0018] Fig. 5 is an MSK autocorrelation function curve with digital sampling points for 0,

-0.25, and 0.5 timing offset with a sampling rate of 1, which corresponds to a sampling rate equal to the chip or symbol rate;

[0019] Fig. 6 is an MSK autocorrelation function curve with digital sampling points for -

0.1 timing offset with a sampling rate of 2, which corresponds to a sampling rate equal to two times the chip or symbol rate;

[0020] Fig. 7 is a conceptual block diagram illustrating a specific example of a system with ^XA clock accuracy using receiver clock time alignment by selecting the matched filter coefficients based on a selection algorithm, which uses the correlation curve according to an embodiment of the present invention; and

[0021] Fig. 8 is a modification to the conceptual block diagram of Fig. 7 for coherent demodulation.

[0022] In the figures, it will be understood that like numerals refer to like features and structures.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS [0023] Fig. 1 is a conceptual block diagram illustrating an exemplary system 100 that can be employed in a receiver to perform receiver clock time alignment with the receive signal according to an embodiment of the present invention. For example, system 100 can be employed in a receiver of user terminals, fixed routers and access points of an ad-hoc network as described in U.S. Patent Application Serial No. 09/897,790 entitled "Ad Hoc Peer-to-Peer Mobile Radio Access System Interfaced to the PSTN and Cellular Networks", filed on June 29, 2001, and in U.S. Patent Application Serial No. 09/815,157 entitled "Time Division Protocol for an Ad-Hoc, Peer-to-Peer Radio Network Having Coordinating Channel Access to Shared Parallel Data Channels with Separate Reservation Channel", filed on March 22, 2001, the entire content of both of said patent applications being incorporated herein by reference in their entirety. Also, System 100 can also be employed in a system using concatenated spreading codes as described in pending patent application Serial No. 10/053,695 entitled "A System and Method Employing Concatenated Spreading Sequences to Provide Data Modulated Spread Signals Having Increased Data Rates with Extended Multi-Path Delay Spread", filed on January 24, 2002, the entire contents being incorporated herein by reference.

[0024] As shown in Figure 1, the system 100 includes a Synchronization and Serial Probe (SYNC/SP) matched filter 102 which filters the received inphase (I) & quadrature (Q) signals during the initial receiver synchronization and subsequent serial probe sections of the received waveform to achieve and maintain timing alignment. Filter coefficients for the SYNC/SP matched filter are selected to receive an on-time signal. The filter coefficients for the SYNC/SP matched filter are not adjusted during either the SYNC or serial probe section of the waveform in order to provide a common reference point for selection of the Rake Taps. [0025] The filtered signal is processed by the SYNC phase rotator 104, thus enabling serial demodulation to be used for modulation waveforms like Minimum Shift Keying (MSK). Serial demodulation offers hardware advantages over a parallel demodulation architecture as discussed in a publication by F. Amoroso and J. Kivett entitled "Simplified MSK Signaling Technique," IEEE Trans. Commun., Vol. COM-25, pp. 433-441, April 1997; in a publication by D. J. Rasmussen and G. Davis entitled "Serial Demodulation of an OQPSK Direct Sequence Spread Signal," IEEE Proceedings of the Tactical Communications Conference, pp. 171-179, May 1992; and in a publication by D. J. Rasmussen entitled "Serial Demodulation of Offset Quadrature Pulse-Shaped Signals," PhD Dissertation, Arizona State University, Tempe Arizona, May 1993, each of which is incorporated herein by reference in its entirety. Serial demodulation structure enables not only MSK modulation waveform to be used, but all the Offset Quadrature Pulse-Shaped Signals defined by general modulation equations 14-a and 14-b in the publication by D. J. Rasmussen entitled "Serial Demodulation of Offset Quadrature Pulse-Shaped Signals," PhD Dissertation, Arizona State University, Tempe Arizona, May 1993, the entire contents of which is incorporated herein by reference. These equations are as follows: M s(t) = ∑dd_22kk ff((kk))qq((tt --22kkTT))ccooss((22^^ ff₀₀tt))++££d_2k+1 f(k) q(t -[2k + l]r)sin(2^ f₀t) k=0 k=0

and

f(k) = (-l)^k ;

where s(t) is the modulated signal, d_2k and d_{2k +}ι represents the transmitted data bits, q(t) is the pulse-shaping function, T is the symbol period, 2M+2 is the total number of transmitted data bits, and f₀ is the carrier frequency. This signal structure is easily expanded to a direct sequence spread spectrum signal by modifying these equations to the following:

M (k+l)N-l

∑c_2n f(n) q(t - 2nT_c)cos(2 f₀t + 0_k) k=0 n=kN

and

f(n) =(-i)ⁿ ;

where s(t) is the modulated signal, θ represents the data phase symbol (0 or π for BPSK or DBPSK data modulation and 0, π/2, π, -π/2 for QPSK or DQPSK data modulation), q(t) is the chip pulse-shaping function, T_c is the chip period, N is the number of chips per data symbol, M+l is the total number of transmitted data symbols, and f₀ is the carrier frequency. These equations assume the same spreading sequence is used for each data symbol. A general equation which does not limited the same spreading sequence to each data symbol is given by the following equation:

^ )];

where the equation for f(n) is the same as defined previously and the INT function takes an integer value of the argument. Such specific modulation waveforms presented in this PhD Dissertation are Serial Offset Quadrature Phase Shift Keying (SOQPSK), Serial Raised Cosine OQPSK (SRC-OQPSK), and Serial Quasi-Bandlimited Minimum Shift Keying (SQBL-MSK). Using the general equations for Offset Quadrature Pulse-Shaped Signals, serial OQPSK using more spectral efficient pulse-shaping like Nyquist pulse-shaping with Raised Cosine Spectrum and pulse-shaping with Square-Root Raised Cosine Spectrum are also included in the modulation signals that are serial demodulated.

[0026] If a parallel MSK demodulation architecture is used, the SYNC phase rotator is removed. The SYNC phase rotator is also not necessary for the demodulation of a Binary Shift Keyed (BPSK) signal or a Quadrature Phase Shift Keyed (QPSK) signal. A QPSK signal requires a parallel demodulation architecture, which like parallel MSK demodulation does not require the SYNC phase rotator. The serial inphase (I) and quadrature (Q) signals out of the SYNC phase rotator 104 are used to determine receiver initial synchronization and the coarse timing control for the Rake processor 118 in the SYNC/SP processing unit 106. The receiver SYNC and frame SYNC used for coarse timing control can be obtained from the same section or separate sections of the signal waveform structure. As can readily appreciated by one skilled in the art, either approach can be used within system 100. [0027] To illustrate the coarse timing resolution provided by SYNC/SP processing unit 106, an example framing sequence consisting of 1023 chips or symbols derived from a maximal-length PN generator will now be described. The sliding correlation operation between the received code and the receiver reference code is shown in Fig. 2 for a coherent receiver with ideal timing and a sampling clock equal to the chip or symbol rate of the received signal. The peak in the correlation function is used to establish frame sync. A received signal level of zero was assumed before the framing sequence with the framing sequence followed by the same 1023 chip or symbol PN sequence. Additional variance in the correlation function will be introduced by receiver noise and any SYNC sequence preceding the framing sequence. Preferably, a SYNC sequence with good cross-correlation properties with the frame sequence is selected. Also, the frame sequence provides reliable detection over the signal-to-noise ratio levels expected over the operating range of the system. As will be appreciated by one skilled in the art, a non-coherent architecture can be used to achieve similar correlation results.

[0028] As discussed above, the peak in the correlation function is used to established frame sync. By observing the correlation function over a selected multi-path window, one is able to determine the time location of the Rake taps for a Rake receiver. Fig. 3 illustrates an example of a sliding code correlation function for equal amplitude direct and multi-path signal into the receiver. The multi-path signal is delayed 40 time samples (chips or symbols) from the direct path. Selection of the two Rake taps is based on the two correlation peaks. Since the channel conditions are changing especially for a mobile communications, serial probes are typically periodically inserted into the waveform to re-establish the channel conditions and select the proper Rake taps.

[0029] Referring again to Fig. 1, the SYNC/SP processing unit 106 is used to establish coarse timing. The coarse timing control taps 1 through N are used to select the proper aligning in the N-Tap Rake Combiner and Demodulator 118. For a direct sequence system, the coarse timing control taps 1 through N select the proper receiver reference spreading code delays so the different delayed received multi-path signals are despread with the time aligned spreading code. Fig. 2 and 3 showed ideal timing conditions where this receiver section is able to obtain ideal timing, which may not be practical under real conditions, but is being introduced in order to present a simple example on how the algorithm works. Also, the sliding correlation functions did not include the transmit pulse-shaping function and receive matched filter, which spreads the correlation peaks shown in Fig. 2 and 3. For the receiver matched to the transmit pulse-shape, the convolution operation in the receiver results in the autocorrelation function or squared autocorrelation function of the modulation signal depending on the receiver architecture. A coherent design shall result in the autocorrelation function, while different non-coherent design will provide either the magnitude or squared autocorrelation function. Since the autocorrelation function is a positive value function, the magnitude autocorrelation function is equal to the autocorrelation function. [0030] Autocorrelation functions for BPSK and MSK example modulation signals are given in Fig. 4 along with the squared autocorrelation functions. Similar autocorrelation and squared autocorrelation functions directly related to the modulation pulse-shape and demodulator matched filtering result for pulse-shaped QPSK and Offset Quadrature Pulse- Shaped signals. Spreading of the correlation peaks shown in Fig. 2 and 3 results from the autocorrelation function. Non-peak levels of the sliding code correlation function are also impacted by the autocorrelation function, but their low signal levels results in them not being used in the timing control selection. , Fig. 4 shows that the autocorrelation function provides larger spreading on the correlation peaks than the squared autocorrelation function. Also, the more bandwidth efficient MSK modulation signal provides larger spread on the correlation function. Spectral efficient modulation waveforms like serial OQPSK with Nyquist pulse- shaping for Raised Cosine Spectrum and pulse-shaping for Square-Root Raised Cosine Spectrum shall provide a larger spread on the correlation function due to the larger pulse- shaping time response.

[0031] The autocorrelation and squared autocorrelation functions shown in Fig. 4 are presented as continuous time signals versus discrete time samples in order to show the desired function to be fit in order to provide fine timing alignment by proper control of the matched filter coefficients. Digital time samples out of the SYNC/SP processing unit 106 are sent to the Correlation Function Comparison unit 108 to determine the fine timing control for taps 1 through N. Applying the proper sampling rate and timing offset on the continuous time curves shown in Fig. 4 resuljts in the digital time samples sent to the Correlation Function Comparison unit 108.

[0032] For example, the digital samples for the MSK autocorrelation function for timing offsets of 0, -0.25, and 0.5 with a sampling rate of 1 (sampling rate equal to the signal chip or symbol rate) are given in Fig. 5. For this specific sampling rate, timing offsets of 0 and -0.25 have three signal conditions greater than 10 % of the on-time peak value. The 0.5 timing offset has two values that are greater than the 10 % of the on-time peak value and two below this level. For the 0.5 timing offset condition, the coarse timing selection determines whether it is a positive or negative timing offset. A coarse timing reference at the -0.5 timing point corresponds to a -0.5 timing offset, while a coarse timing reference at the 0.5 timing point corresponds to a 0.5 timing offset.

[0033] Increasing the sampling rate from 1 to 2, with a -0.1 timing offset, results in the samples of the MSK autocorrelation function as shown in Fig. 6. The increased sampling rate provides 5 samples greater than 10 % of the on-time peak for this timing offset condition. Using a higher sampling rate will increase the number of samples available for the curve fitting operation to determine the fine timing control for taps 1 through N at the expense of increased hardware and clock speed.

[0034] The fine time alignment for each Rake tap is determined by curve fitting the digital samples to the known autocorrelation or squared autocorrelation function. Different curve fitting techniques readily appreciated by one skilled in the art can be used within the correlation function comparison unit 108 to achieve the fine timing control. As shown, the number of samples for curve fitting is improved by increasing the sampling rate. Instead of burdening the entire receiver architecture with a higher clock rate, an interpolation operation can be performed within the Correlation Function Comparison unit 108 for providing increased samples for the fine timing estimation.

[0035] The Correlation Function Comparison unit 108 establishes the fine timing control requirement for each of the Rake taps 1 through N. Applying the desired fine timing control for each Rake tap is achieved by proper selection of the matched filter coefficients for each Rake tap. The received signal for each identified Rake tap is aligned with the appropriate matched filter response. The filter coefficient array based on timing resolution 110 stores all the available matched filter coefficients available for fine time alignment. System timing requirements will determine the size requirement of the filter coefficient array unit 110. [0036] It will be readily appreciated that finer timing accuracy can be achieved with a larger filter coefficient array unit 110. Multiplexers 112-1 through 112-N are controlled by the appropriate fine timing control signals from Correlation Function Comparison Unit 108 to load the matched filters 114-1 through 114-N with the appropriate filter coefficients from the filter coefficient array 110.

[0037] To demonstrate how this filter coefficient selection process works, the samples for the digitized MSK autocorrelation function shown in Fig. 5 is used. Assume that the timing offset of -0.25, 0, and +0.5 corresponds to Rake tap 1 through 3, respectively. The filter coefficients loaded into matched filter 114-1 will consist of the matched filter with a -0.25 timing offset. Matched filter 114-2 will be loaded with on-time matched filter coefficients. Matched filter 114-3 will be loaded with matched filter coefficients with a +0.5 timing offset. [0038] Assuming the SYNC/SP Processing 106 has not detected any signals to identify with Rake taps 4 through N, the matched filter 114-4 through 114-N will not be adjusted. Therefore, the output of matched filter 114-4 through 114-N will not be used in the N-Tap Rake Combiner and Demodulator 118, since no Rake taps for these elements have been detected. By applying the proper timing offsets to the matched filters, the received signal for each of the Rake tap is aligned properly with the matched filter operation maximizing the receiver signal-to-noise ratio under a white Gaussian noise condition. [0039] The output of the matched filter 114-1 is applied to phase rotator 116-1, matched filter 114-2 output is applied to phase rotator 116-2, and so on for matched filter 114-3 through 114-N and phase rotator 116-3 through 116-N. Phase rotators 116-1 through 116-N are used to perform serial demodulation on signals such as MSK and other Offset Quadrature Pulse-Shaped signals. For BPSK and QPSK signals, phase rotators 116-1 through 116-N are removed, and the outputs of the matched filters 114-1 through 114-N are directly connected to the appropriate tap input of the N-Tap Rake Combiner and Demodulator 118. For serial demodulation, phase rotators 116-1 through 116-N are connected to the appropriate tap input of the N-Tap Rake Combiner and Demodulator 118. The N-Tap Rake Combiner and Demodulator 118 performs the despreading operation, Rake symbol combining, and symbol demodulation for a direct sequence signal. For a non-spread signal, N-Tap Rake Combiner and Demodulator 118 operates directly at the symbol rate rather than at a higher chip rate. Since the non-spread signal does not have the spreading code properties to isolate the multi- path signals, the Rake combiner is replaced with a channel equalizer. It will be readily appreciated by those of skill in the art that the channel equalizer reduces inter-symbol interference by canceling out the channel effects. Typically, a training sequence is added to the waveform in order to estimate the amplitude and phase parameters for each multi-path tap identified by the SYNC/SP Processing Unit 106.

[0040] As discussed in connection with the exemplary embodiments of the present invention, the timing alignment can be applied to a direct sequence spread spectrum or non- spread communication system. Since a direct sequence spread spectrum spreads the symbol information over many chips, the chip timing requirements are typically less stringent. The less stringent chip-timing requirement enables the hardware complexity to be reduced for the higher speed chip processing. For direct sequence spread spectrum systems with chip timing resolution of ± 0.25 of a chip period and the demodulator sampling rate equal to the chip rate, the simplified receiver system 200 shown in Fig. 7 can be used. The receiver in Fig. 7 also assumes a non-coherent carrier phase recovery architecture is used. This assumptions enables one phase rotator to be used for both the -0.5 and 0.5 chip period (Tc) delayed signals. Using the same phase rotator results in different phase conditions on the signals. For a non-coherent system, this phase shall be removed with a differential detector. For a coherent carrier architecture, the correct phase on each of the delayed signals is required. An extra phase rotator as shown in Fig. 8 is required to generate the appropriate signals to the fine timing control multiplexers.

[0041] Comparing receiver system 200 to receiver system 100, the hardware is significantly simplified by the reduction of the number of chip matched filters and phase rotators along with the removal of the filter coefficient array. Receiver system 200 uses only two chip matched filter structures. Chip matched filter 202 uses on-time filter coefficients and combined with phase rotator 204 provides the on-time detected signal for the SYNC Processor 106 and N-Tap Rake Combiner and Demodulator 118. Chip matched filter 210 uses -0.5 Tc delayed filter coefficients, which are the same filter coefficients as that required for 0.5 Tc delay. Phase rotator 212 follows the chip matched filter with -0.5 Tc filter coefficients to provide the -0.5 Tc delayed detected signal for the N-Tap Rake Combiner and Demodulator. For a timing offset of 0.5 Tc, the signal out of the chip matched filter with - 0.5 Tc filter coefficients shall be a chip period (Tc) ahead of the defined peak location defined by the coarse timing control. A single chip delay element (1 Chip Delay) 214 is applied to the output of phase rotator 212 to place the chip matched filter output at the appropriate sample location. The output of the 1 chip delay 214 provides the 0.5 Tc delayed detected signal for the N-Tap Rake Combiner and Demodulator.

[0042] As previously discussed, the differential detection process enables only one phase rotator to be used for the generation of the -0.5 and 0.5 Tc delayed detected signals. The 1 chip delay 214 required for the 0.5 Tc delayed detected signal can be placed after the phase rotator since the differential detection does not require an exact phase for the phase rotator. As with receiver system 100, receiver system 200 requires the phase rotator for serial demodulation of spreading signals like MSK and other Offset Quadrature Pulse-Shaped signals. For reception of BPSK and QPSK spreading signals, the phase rotators 204 and 212 can be removed.

[0043] The on-time detected signal out of phase rotator 204 is sent to the SYNC/SP Processing 206. As discussed with receiver system 100, receiver system 200 determines coarse timing for each of the Rake Taps in the SYNC/SP Processing 206. A known frame sequence is used to obtain correlation peaks to identify the Rake taps. The multi-path window determines the range over which the Rake taps can be established. For the direct sequence spread system 200, the coarse timing control taps 1 through N select the proper receiver reference spreading code delays so the different delayed received multi-path signals are despread with the time aligned spreading code. For mobile communications, serial probes are typically inserted periodically into the waveform to re-establish the channel conditions and select the proper Rake taps.

[0044] The sliding correlation signal out of the SYNC/SP Processing Unit 206 is sent to the Correlation Function Comparison Unit 208 to establish the fine timing control for Rake tap 1 though N. As with receiver system 100, receiver system 200 obtains the fine time alignment for each Rake tap by curve fitting the digital samples from SYNC/SP Processing Unit 208 to a known autocorrelation or squared autocorrelation function. It will be readily appreciated by one skilled in the art that different curve fitting techniques can be used within the correlation function comparison unit 108 to achieve fine timing control. Since only three different timing conditions exist by minimizing the number of chip matched filter time offsets to three, the Correlation Function Comparison Unit 208 needs to map the estimated time offset to one of these fine timing conditions. One such mapping, maps the timing offset (To) as follows:

1) -0.25 T_c < T₀ < 0.25 T_c mapped to on-time detected signal.

2) -0.5 T_c < T₀ < 0.25 T_c mapped to -0.5 T_c detected signal.

3) 0.25 To < T_o < 0.5 T_c mapped to 0.5 T_c detected signal.

[0045] As will be appreciated by one skilled in the art, different mapping conditions can be implemented to meet the design requirements of the particular system being designed. Based on the mapping algorithm the fine timing control for tap 1 controls the output of multiplexer 216-1, fine timing control for tap 2 controls the output of multiplexer 216-2, and so on for the remaining multiplexers. Muliplexers 216-1 through 216-N are connected to the Rake tap inputs 1 through N of the N-Tap Rake Combiner and Demodulator 218, respectively. The N-Tap Rake Combiner and Demodulator 218 performs the despreading operation, differential signal detection for each Rake tap, the Rake symbol combining, and symbol detection. [0046] Although only a few exemplary embodiments of the present invention have been described in detail above, those skilled in the art will readily appreciate that many modifications and variations are possible in the exemplary embodiments without materially departing from the novel teachings and advantages of this invention. Accordingly, all such modifications are intended to be included within the scope of this invention as defined in the following claims.

Claims

What is claimed is:

1. A system for establishing timing alignment between an input signal and reference signals of a receiver in a communications network, the system comprising: a coarse synchronization unit, adapted to compare said input signal to a synchronization pattern and to generate a coarse timing signal, said coarse timing signal being adapted to set the coarse timing of a demodulator, and said coarse synchronization unit being further adapted to output a correlation signal; a correlation function comparison unit, adapted to compare said correlation signal to an expected autocorrelation curve, and based on said comparison, to generate a fine timing correction signal; and a programmable matched filter, adapted to select a coefficient of delay based on said fine timing correction signal, to delay said input signal to substantially synchronize said input signal to said timing of said demodulator.

2. A system as claimed in claim 1, further comprising: a multiplexer, adapted to select one of a plurality of multiplexer inputs to connect to a multiplexer output based on said fine timing correction signal, said multiplexer output controlling said programmable matched filter to select said coefficient of delay.

3. A system as claimed in claim 2, further comprising: a filter coefficient array having a plurality of vector outputs, each of said vector outputs connected to the multiplexer inputs, each of said vector outputs corresponding to one of a plurality of possible coefficients of delay.

4. A system as claimed in claim 1, wherein said demodulator comprises a plurality of rake taps adapted to compensate for multipath effects, and wherein said coarse synchronization unit is adapted to generate a plurality of coarse timing signals corresponding to each rake tap.

5. A system as claimed in claim 4, further comprising a plurality of programmable matched filters corresponding to each of the rake taps, and wherein said correlation function comparison unit is adapted to generate a plurality of fine timing correction signals corresponding to each of said programmable matched filters.

6. A system as claimed in claim 2, further comprising a plurality of said multiplexers corresponding to each of said plurality of programmable matched filters, and each of said plurality of fine timing correction signals.

7. A system as claimed in claim 1, wherein said correlation signal comprises a plurality of digital samples.

8. A system as claimed in claim 1, wherein said input signal includes a BPSK signal.

9. A system as claimed in claim 1, wherein said input signal includes an MSK signal.

10. A system as claimed in claim 1, wherein said input signal includes a QPSK signal.

11. A system as claimed in claim 1, wherein said input signal includes an OQPSK signal.

12. A system as claimed in claim 1, wherein said coarse timing signal synchronizes said input signal to said demodulator within a single clock period.

13. A system as claimed in claim 1, wherein said fine timing signal synchronizes said input signal to said receiver within a quarter clock period.

14. A system as claimed in claim 1, wherein said fine timing signal synchronizes said input signal to said receiver within less than a quarter clock period.

15. A system as claimed in claim 1, wherein said correlation function comparison unit compares said correlation signal samples to an expected squared autocorrelation function.

16. A method of controlling a receiver in a communication network to synchronize an input signal to reference signals of the receiver, comprising: receiving said input signal; comparing said input signal to a signal pattern; based on said comparison, generating a coarse timing control signal adapted to substantially synchronize a demodulator to within a clock period of said input signal, and to generate a correlation signal; comparing said correlation signal to an expected autocorrelation, and based on said comparison, generating a fine timing control signal; based on said fine timing control signal, controlling a programmable matched filter to select a coefficient of delay, said coefficient determining a delay to be applied to said input signal, thereby substantially synchronizing said demodulator to said delayed input signal.

17. A method as claimed in claim 16, further comprising: applying said fine timing control signal to at least one multiplexer, said fine timing control signal controlling said multiplexer to connect one of a plurality of multiplexer inputs to said programmable matched filter, and thereby controlling the programmable matched filter to select a coefficient of delay.

18. A method as claimed in claim 17, wherein said multiplexer inputs are connected to a filter coefficient array, said filter coefficient array comprising a plurality of vector signals adapted to control said programmable matched filters to select a coefficient of delay.

19. A method as claimed in claim 16, wherein said demodulator comprises a plurality of rake taps adapted to compensate for multipath effects, and wherein said generating a coarse timing signal step further comprises generating a plurality of coarse timing signals corresponding to each rake tap.

20. A method as claimed in claim 19, further comprising a plurality of programmable matched filters, and wherein the step of generating a fine timing control signal further comprises generating a plurality of fine timing control signals corresponding to each of said plurality of programmable matched filters, said plurality of programmable matched filters corresponding to each of said rake taps.

20. A method as claimed in claim 17, further comprising a plurality of said multiplexers corresponding to each of said plurality of programmable matched filters, wherein said step of generating a fine timing control signal further comprises generating a plurality of fine timing control signals corresponding to each of said multiplexers, programmable matched filters and rake taps.

22. A method as claimed in claim 16, wherein said correlation signal comprises a plurality of digital samples.

23. A method as claimed in claim 16, wherein said input signal includes a BPSK signal.

24. A method as claimed in claim 16, wherein said input signal includes an MSK signal.

25. A method as claimed in claim 16, wherein said input signal includes a QPSK signal.

26. A method as claimed in claim 16, wherein said input signal includes an OQPSK signal.

27. A method as claimed in claim 16, wherein said controlling a programmable matched filter step further comprises synchronizing said demodulator to within a quarter clock period of said delayed input signal.

26. A method as claimed in claim 16, wherein said comparing said correlation signal step further comprises comparing said correlation signal to an expected squared autocorrelation curve.

29. A computer readable medium of instructions adapted to control a receiver in a communications system to synchronize to a received signal comprising: a first set of instruction adapted to control the receiver system to receive said input signal, a second set of instructions adapted to control the receiver to compare said input signal to a signal pattern, a third set of instructions adapted to control the receiver system to generate a coarse timing signal based on said comparison, in order to synchronize a demodulator to within a clock period of said input signal, and to generate a correlation signal based on said comparison, a fourth set of instructions adapted to control the receiver system to compare said correlation signal to an expected autocorrelation curve and, based on said comparison, to generate a fine timing control signal, and a fifth set of instructions, adapted to control said receiver to control a programmable matched filter based on said fine timing control signal to select a coefficient of delay, said coefficient determining a delay to be applied to said input signal, thereby substantially synchronizing said demodulator to said delayed input signal.

30. A computer readable medium of instructions of claim 29, wherein the fifth set of instructions is adapted to control at least one multiplexer based on said fine timing control signal to connect one of a plurality of multiplexer inputs to said programmable matched filter, thereby controlling the programmable matched filter to select a coefficient of delay.

31. A computer readable medium of instructions of claim 30, wherein a plurality of vector signals from a filter coefficient array are connected to said multiplexer inputs, and wherein said fifth set of instructions is further adapted to control said multiplexer based on said fine timing control signal to connect one of said vector signals to said programmable matched filter to select a coefficient of delay.

32. A computer readable medium of instructions of claim 29, wherein said demodulator comprises a plurality of rake taps adapted to compensate for multipath effects, and wherein said third set of instructions is adapted to generate a plurality of coarse timing signals corresponding to each of said rake taps.

33. A computer readable medium of instructions of claim 29, wherein said fifth set of instructions is further adapted to control a plurality of programmable matched filters based on a corresponding plurality of fine timing control signals.

34. A computer readable medium of instructions of claim 30, wherein said fifth set of instructions is adapted to control a plurality of multiplexers based on a corresponding plurality of fine timing control signals.

35. A computer readable medium of instructions of claim 29, wherein said third set of instructions is adapted to generate a con-elation signal comprising a plurality of digital samples.

36. A computer readable medium of instruction of claim 29, wherein said first set of instructions is adapted to control said receiver to receive a BPSK signal.

37. A computer readable medium of instruction of claim 29, wherein said first set of instructions is adapted to control said receiver to receive an MSK signal.

38. A computer readable medium of instruction of claim 29, wherein said first set of instructions is adapted to control said receiver to receive a QPSK signal.

39. A computer readable medium of instruction of claim 29, wherein said first set of instructions is adapted to control said receiver to receive an OQPSK signal.

40. A computer readable medium of instruction of claim 29, wherein said fifth set of instructions is adapted to synchronize said demodulator to within a quarter clock period of said delayed input signal.

41. A computer readable medium of instruction of claim 29, wherein said fourth set of instructions is adapted to control the receiver system to compare said correlation signal to an expected squared autocorrelation curve.