US20040151238A1 - Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth - Google Patents
Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth Download PDFInfo
- Publication number
- US20040151238A1 US20040151238A1 US10/758,355 US75835504A US2004151238A1 US 20040151238 A1 US20040151238 A1 US 20040151238A1 US 75835504 A US75835504 A US 75835504A US 2004151238 A1 US2004151238 A1 US 2004151238A1
- Authority
- US
- United States
- Prior art keywords
- signal
- receiver
- transceiver
- filter
- digital
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Abandoned
Links
Images
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/06—Receivers
- H04B1/10—Means associated with receiver for limiting or suppressing noise or interference
- H04B1/12—Neutralising, balancing, or compensation arrangements
- H04B1/123—Neutralising, balancing, or compensation arrangements using adaptive balancing or compensation means
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/38—Transceivers, i.e. devices in which transmitter and receiver form a structural unit and in which at least one part is used for functions of transmitting and receiving
- H04B1/40—Circuits
- H04B1/50—Circuits using different frequencies for the two directions of communication
- H04B1/52—Hybrid arrangements, i.e. arrangements for transition from single-path two-direction transmission to single-direction transmission on each of two paths or vice versa
- H04B1/525—Hybrid arrangements, i.e. arrangements for transition from single-path two-direction transmission to single-direction transmission on each of two paths or vice versa with means for reducing leakage of transmitter signal into the receiver
Definitions
- the present invention relates to wireless communications. More specifically, the invention relates to canceling transmitter signal energy in a receiver bandwidth.
- CDMA code division multiple access
- GSM global system for mobile communications
- TDMA time division multiple access
- a standard transceiver comprises a transmitter, a receiver and a duplexer.
- One problem with standard transceivers is some of the transmit signal leaks into the receiver, which corrupts the received signal processed by the receiver.
- the duplexer functions to reduce the transmit signal received by the receiver through the use of one or more circulators, receive filters and/or transmit filters.
- a circulator within the duplexer directs the transmit signal toward an antenna radiator, while a receive filter rejects the residual transmit energy directed toward the receiver that falls outside of the receive filter bandwidth.
- some duplexers do not use a circulator.
- Imperfect duplexers create two problems.
- the high-level transmit signal from the transmitter drives the receiver into a non-linear operating region.
- This problem can be solved by increasing the rejection characteristics of the receive filter to more effectively attenuate the main spectral lobe of the transmitter signal.
- this problem may be mitigated through the use of a high dynamic range, direct conversion, digital receiver that can implement the requisite filter characteristics with lossless digital filtering.
- a second problem is noise from the transmitter leaks into the receiver and raises its noise figure.
- the second problem can be mitigated by further filtering the transmitter output and by reducing the spectral sidelobe energy through signal design and linearization of a transmit amplifier. These changes, however, are undesirable from the standpoint of transmitter efficiency.
- the present invention relates to a method and apparatus for adaptive digital cancellation of a transmit signal spectrum in a receiver bandwidth.
- the apparatus relates to a digital coherent spectral canceller that attenuates the spectral components from the transmitter that fall within the bandwidth of the receiver.
- the digital adaptive coherent spectral canceller digitizes both a corrupted receiver signal and a reference transmit signal and then digitally implements an adaptive coherent spectral canceller adaptation module.
- One embodiment of the invention is implemented with a direct conversion digital receiver, which achieves an image-free, high dynamic range without the use of automatic gain control.
- Automatic gain control used to extend the dynamic range of a receiver may be undesirable for spectrally crowded applications such as cellular communications because the automatic gain control may make the receiver sensitivity dependent upon signals and interference that are outside the signal channel. For example, it is possible for a strong signal in an adjacent channel to capture the receiver front end desensitize the receiver such that a weak signal in the channel of interest is undetectable. This is particularly harmful in a base station receiver where the receiver receives incoming signals from multiple remote units.
- the use of automatic gain control will likely require the digital coherent canceller to track the gain changes which will introduce errors and noise.
- One embodiment of the invention uses an adaptive transversal filter in an all digital implementation to compensate for the amplitude and phase differences between the transmitter-to-receiver leakage path and the transmit signal path utilized as a reference over the receiver bandwidth or bandwidth of interest.
- An advantage of the invention is reduced complexity and cost, which is achieved through: (1) reduced performance requirements of the duplexer, (2) reduction in the requirement for transmit signal filtering, and (3) reduction in the requirements for high linearity or linearization of the transmit amplifier.
- the canceller system is configured to attenuate a signal spectrum from a transmitter which falls within a bandwidth of a receiver.
- the canceller system comprises a reference bandpass filter, a reference direct converter, a cross correlation measurer, an adaptation coherent spectral canceller algorithm module executed, for example, on a microcontroller, an adaptive digital transversal filter, and a combiner.
- the reference direct converter is adapted to output a digitized transmit signal reference of a spectral energy of the transmitter within the bandwidth of the receiver.
- the adaptive digital transversal filter is adapted to align an amplitude and phase of a digitized transmit signal reference in a reference path with a transmit signal in a leakage receiver path.
- the adaptive digital transversal filter outputs a compensated or equalized digitized transmit signal reference.
- the combiner is adapted to coherently subtract the compensated, digitized transmit signal reference from a corrupted, digitized receiver signal to form a residue, having transmitter spectral signal power within the bandwidth of a receiver is suppressed.
- Another aspect of the invention relates to a method of attenuating a transmit signal spectrum in a bandwidth of a receiver.
- the method comprises digitizing a received signal which is corrupted by components of a transmit signal, creating a digitized reference transmit signal of the transmit signal within the bandwidth of the receiver, aligning the digitized reference transmit signal in amplitude, phase and time delay with the digitized received signal, subtracting the digitized reference transmit signal from the digitized received signal to form a residue, and suppressing a transmitter spectral signal power of the residue within the bandwidth of the receiver.
- FIG. 1 illustrates one embodiment of a transceiver and a processor in accordance with the present invention.
- FIG. 2A illustrates transmitter spectral spillage into a receiver bandwidth with a single channel.
- FIG. 2B illustrates transmitter spectral spillage into a receiver bandwidth with multiple channels.
- FIG. 3 illustrates one embodiment of a communication system with a transceiver in accordance with the present invention.
- FIG. 4 illustrates an exemplary spectral density at an input and output of an adaptive coherent spectral canceller within the transceiver of FIG. 1.
- FIG. 5A illustrates one embodiment of a transceiver with a shared transmit and receive antenna radiator.
- FIG. 5B illustrates one embodiment of a transceiver with separate receive and transmit antenna radiators.
- FIG. 6 illustrates one embodiment of a digital coherent spectral canceller within the transceiver of FIGS. 5A and 5B.
- FIG. 7 illustrates one embodiment of a direct converter within the transceiver of FIGS. 5A and 5B.
- FIG. 8 illustrates another embodiment of a direct converter within the transceiver of FIGS. 5A and 5B.
- FIG. 9A illustrates exemplary baseband responses of one embodiment of a receive filter and a transmit reference filter within the transceiver of FIG. 5A.
- FIG. 9B illustrates cancellation ratios of spectral leakage from the transmitter as a result of a digital coherent spectral canceller of FIG. 5A having varying number of taps with the filter responses of FIG. 9A.
- FIG. 10A illustrates exemplary baseband responses of another embodiment of the receive filter and transmit reference filter within the transceiver of FIG. 5A.
- FIG. 10B illustrates exemplary outputs of one embodiment of a digital spectral cancellers of FIG. 5A having varying numbers of taps with the filter responses of FIG. 10A.
- FIG. 11 illustrates another embodiment of a direct converter within the transceiver of FIGS. 5A and 5B.
- FIG. 12 illustrates another embodiment of a direct converter within the transceiver of FIGS. 5A and 5B.
- FIG. 13 illustrates one embodiment of a translating delta-sigma modulator within the direct converter of FIGS. 7 and 11.
- FIGS. 14 A- 14 E are spectral plots used to illustrate the operation of various embodiments of the transceiver of FIG. 1.
- FIG. 15 is a block diagram showing one embodiment of a component within FIGS. 11 and 12.
- FIG. 16 is a block diagram showing one embodiment of a clock generator of FIG. 7.
- FIG. 17 illustrates another embodiment of a modulator within the direct converters of FIGS. 7 and 11.
- FIG. 1 illustrates an exemplifying communication system in which the present invention may be implemented. Alternatively, the present invention may be used in other systems and applications.
- FIG. 1 illustrates one embodiment of a transceiver 200 and a processor 900 in accordance with the present invention.
- the transceiver 200 is a full duplex transceiver, although the invention can be used in communication systems in which the base station concurrently transmits and receives regardless of whether the communication system provides full duplex operation.
- the transceiver 200 is for code division multiple access (CDMA) mobile and base station transceivers. In another embodiment, the transceiver 200 is for a time division multiple access (TDMA) transceiver, such as a transceiver used for GSM and IS-54 base stations.
- the transceiver 200 comprises transmit and receive antenna radiator 102 , a duplexer 210 , a receiver 106 , a transmitter 280 and a digital coherent spectral canceller 230 .
- the receiver 106 is a direct conversion digital receiver described herein. In another embodiment, the receiver 106 is a modified direct conversion digital receiver described in U.S. Pat. No. 5,557,642, entitled “Direct Conversion Receiver For Multiple Protocols” by Williams modified to exclude automatic gain control.
- the transmitter 280 and the digital coherent spectral canceller 230 are coupled to the processor 900 .
- the transmit and receive antenna radiator 102 is coupled to the duplexer 210 , which is coupled to the receiver 106 , the digital coherent spectral canceller 230 and the transmitter 280 .
- the processor 900 sends data and control signals to the transmitter 280 , which sends data to the duplexer 210 for transmission via the antenna radiator 102 .
- the duplexer 210 sends a reference transmit signal to the canceller 230 .
- the duplexer 210 also receives data from the radiator 102 and sends data to the receiver 106 , which sends the received data to the canceller 230 .
- the canceller 230 and the processor 900 use the reference transmit signal from the duplexer 210 to cancel noise from the transmitter 280 that leaked into the receiver bandwidth.
- noise from the transmitter 280 that leaks into the receiver 106 can be mitigated by further filtering the transmitter output and by reducing the spectral sidelobe energy through signal design and linearization of a transmit amplifier. These changes, however, are undesirable from the standpoint of transmitter efficiency.
- the digital adaptive coherent spectral canceller 230 of the present invention attenuates the spectral components from the transmitter 280 that fall within the bandwidth of the receiver 106 .
- the digital canceller 230 digitizes both a corrupted receiver signal and a reference transmit signal and then digitally implements an adaptive coherent spectral canceller adaptation module with the processor 900 .
- the receiver 106 is preferably a direct conversion digital receiver that achieves an image-free, high dynamic range without the use of automatic gain control.
- Automatic gain control used to extend the dynamic range of a receiver may be undesirable for spectrally crowded applications such as cellular communications because the automatic gain control may make the receiver sensitivity dependent upon signals and interference that are outside the signal channel.
- the digital coherent spectral canceller 230 of the present invention and the methods of using the canceller 230 provide several advantages over conventional systems. These advantages include (1) reducing performance requirements of the duplexer 210 , (2) reducing the requirement for transmit signal filtering, and (3) reducing in the requirements for high linearity or linearization of the transmit amplifier.
- FIG. 2A illustrates a single-channel system with a transmitted signal power spectrum (spectral density) 400 , denoted as ⁇ overscore (S) ⁇ T (f) and centered about a transmit frequency f T , and a received communication signal power 404 , denoted as ⁇ overscore (S) ⁇ R (f) and centered about a receive frequency f R .
- spectral density 400 denoted as ⁇ overscore (S) ⁇ T (f) and centered about a transmit frequency f T
- a received communication signal power 404 denoted as ⁇ overscore (S) ⁇ R (f) and centered about a receive frequency f R .
- the horizontal axis represents frequency (f), such as in units of Gigahertz
- the vertical axis represents signal power amplitude, such as in units of decibels. The horizontal axis has been segmented so that more of the signal energy can be shown.
- FIG. 2A also illustrates a receiver bandwidth 402 from a receiver bandpass filter, such as the receiver bandpass filter 214 described below with reference to FIG. 5A.
- the receiver bandwidth 402 is designed to pass the received signal 404 but not the main transmit signal 400 .
- the transmitted signal power spectrum 400 and the receiver bandwidth 402 are shown in dashed lines.
- FIG. 2A also shows transmit spectral spillage 406 in the receiver bandwidth 402 from a single channel. In other words, noise 406 from the transmitter 280 (FIG. 1) leaks (spills) into the receiver 106 and raises its noise figure.
- the canceller 230 of the FIG. 1 is designed to cancel this transmit noise 406 within the receiver bandwidth 402 .
- FIG. 2B illustrates a multiple-channel system with a plurality of transmitted signal powers 404 ′, 404 ′′, 404 ′′′, a receiver bandwidth 402 and a plurality of received communication signal powers 404 ′, 404 ′′, 404 ′′′.
- FIG. 2B also shows transmitter spectral spillage 406 in the receiver bandwidth 402 from multiple channels (multiple transmit signals and multiple receive signals).
- the plots represent signals which have passed through an amplifier in the receiver 106 of FIG. 1, such as the low noise amplifier 216 shown in FIG. 5A.
- FIG. 4 illustrates an exemplary spectral density at an input and output of the adaptive coherent spectral canceller 230 , as shown in FIG. 1 and described in greater detail below with reference to FIGS. 5A, 5B and 6 .
- FIG. 4 is based on the plots within the receiver bandwidth 402 of FIG. 2B.
- the bandpass transfer function 402 of the receiver bandpass filter 214 passes the three received signal power densities 404 ′, 404 ′′, 404 ′′′ and the transmit noise spectral density 406 .
- a direct converter filter such as the direct converter 220 described below with reference to FIG. 5A, has an even more selective transfer function 408 than the receiver bandpass filter transfer function 402 .
- the direct converter filter transfer function 408 passes only two 404 ′, 404 ′′ of the three signal spectral densities 404 ′, 404 ′′, 404 ′′′ and a portion of the transmit noise spectral density 406 .
- only two received spectral densities 404 ′, 404 ′′ are efficiently passed through the direct converter 220 and on to the canceller 230 (FIGS. 1 and 5A).
- the canceller 230 reduces the transmit noise spectral density 406 to a residual spectral density 410 or lower.
- FIG. 3 illustrates some of the inputs, outputs, signal flow and functions of the processor 900 and the transceiver 200 of FIG. 1.
- FIG. 3 illustrates an antenna radiator 102 , a transceiver 200 , a processor 900 and a transmitter frequency synthesizer and VCO 600 .
- the transceiver 200 comprises an adaptive digital canceller 230 , as shown in FIGS. 1, 5A, 5 B and 6 , in accordance with the present invention.
- the processor 900 comprises a canceller adaptation algorithm module 910 , a digital demodulator module, a frequency control module and a signal waveform clock module.
- the modules may comprise software, firmware, hardware or any combination thereof or may be implemented with discrete logic or in an application specific integrated circuit (ASIC).
- the canceller adaptation module 910 may be executed by a dedicated element, which is separate from the processor 900 .
- the processor 900 comprises a collection of processing elements such as an ARM or a Power Point microprocessor and a dedicated digital modem such as the Qualcomm MSM 3000 .
- the configurations of the embodiments, the configuration of the processor 900 may be different.
- the antenna radiator 102 is coupled to the transceiver 200 , which is coupled to the processor 900 .
- the processor 900 is coupled to the transmitter frequency synthesizer and VCO 600 , which is coupled to the transceiver 200 .
- the processor 900 may be coupled to other components that transfer data and/or control signals to the processor 900 or receive signals from the processor 900 .
- the transceiver 200 transmits and receives signals via the antenna radiator 102 .
- the transceiver 200 passes various data to the processor 900 , such as an I & Q filtered signal and correlation measurement data, C(j), as described below with reference to FIG. 5.
- the underlining of a variable, such as C(j) denotes that the variable is a complex variable, i.e., the variable consists of an in-phase (I) component and a quadrature component (Q).
- the processor 900 may pass various data to the transceiver 200 , such as a clock waveform, CLK, canceller parameters, W(j), and transmitter data.
- the processor 900 may also provide control information, such as a receiver frequency control signal, to the transceiver 200 .
- the processor 900 may also provide control information, such as a transmitter or calibration frequency control signal, to the transmitter frequency synthesizer and VCO 600 , which transmits a signal to the transceiver 200 .
- FIG. 5A illustrates one embodiment of the transceiver 200 of FIG. 3 with a shared receiver/transmitter (Rx/Tx) antenna radiator 102 and an adaptive, digital, coherent spectral canceller 230 .
- the transceiver 200 comprises a duplexer 210 , a low noise amplifier (LNA) 216 , a receiver direct converter 220 , a digital adaptive coherent spectral canceller 230 , a matched or channel filter 250 and a transmitter 280 .
- the duplexer 210 comprises a circulator 212 , a receiver bandpass filter 214 , a transmitter bandpass filter 218 and a directional coupler 235 .
- the canceller 230 comprises a combiner 232 , an adaptive digital transversal filter (or equalizer) 234 , a reference bandpass filter 236 , a reference direct converter 238 and a cross correlation measurement module 240 .
- the duplexer 210 does not comprise a directional coupler 235 , and instead some of the energy passed to the antenna radiator 102 is coupled to an input of the reference bandpass filter 236 within the canceller 230 .
- the transmitter 280 receives data and control signals from a digital processor interface.
- the transmitter 280 is coupled to the transmitter bandpass filter 218 , which is coupled to the directional coupler 235 .
- the directional coupler 235 is coupled to the circulator 212 and the reference bandpass filter 236 .
- the circulator 212 is coupled to the antenna radiator 102 and the receiver bandpass filter 214 , which is coupled to the LNA 216 .
- the LNA 216 is coupled to the receiver direct converter 220 , which is coupled to the combiner 232 , the reference direct converter 238 and the digital processor interface.
- the combiner 232 is coupled to the matched or channel filter 250 and the adaptive digital transversal filter 234 .
- the combiner 232 is also coupled to the cross correlation measurement module 240 .
- the matched filter 250 is coupled to the digital processor interface and the cross correlation measurement module 240 .
- the reference bandpass filter 236 is coupled to the reference direct converter 238 , which is coupled to the receiver direct converter 220 , the digital processor interface, the adaptive digital transversal filter 234 and the cross correlation measurement module 240 .
- Both the adaptive digital transversal filter 234 and the cross correlation measurement module 240 are coupled to the canceller adaptation module 910 , which is executed by the digital processor 900 (FIG. 3) or by a separate microcontroller.
- the antenna radiator 102 receives an incoming signal and transfers the incoming signal to the circulator 212 .
- the incoming signal is a high frequency signal, such as a digitally modulated RF signal centered about a carrier frequency of approximately 2 Gigahertz (GHz).
- GHz Gigahertz
- other signals and carrier frequencies may be used in accordance with the invention.
- the circulator 212 transfers the received signal to the receiver bandpass filter 214 .
- the transmitter 280 may be transmitting a signal to the transmitter bandpass filter 218 , which outputs a transmitter signal S TX (t), to the circulator 212 .
- the signals described herein may be represented as either functions of time, (t), or alternatively, as their Fourier equivalents, functions of frequency, ( ⁇ ).
- the circulator 212 directs the transmit signal to the antenna radiator 102 .
- some of the energy associated with the transmit signal, S TX (t) leaks into the receiver bandpass filter 214 because the circulator 212 does not provide perfect isolation. This leakage signal energy 406 is limited by the receiver bandpass filter 214 , as shown in FIGS. 2A, 2B and 4 .
- the resultant output of the receiver bandpass filter 214 due to S TX (t) and the received signal is input to the LNA 216 .
- the LNA 216 amplifies the signal and passes the signal to the receiver direct converter 220 .
- the receiver direct converter 220 converts the filtered signal 217 to a baseband, digitized and filtered signal to produce an output, X(n).
- the digital coherent spectral canceller 230 adaptively suppresses the transmitter spectral spillage 406 in the receiver band 402 shown in FIG. 4.
- the directional coupler 235 provides a reference signal to the digital coherent spectral canceller 230 .
- the reference signal represents the transmit signal, S TX (t).
- the reference signal is passed to the reference bandpass filter 236 .
- the transfer function H REF ( ⁇ ), of the reference bandpass filter 236 preferably matches the transfer function H RX ( ⁇ ), of the receiver bandpass filter 214 in the previously described leakage path to the receiver direct converter 220 .
- one of the filters 214 , 236 may be designed to cancel a smaller bandwidth (more rejection) compared to the other filter 214 , 236 .
- the reference bandpass filter 236 outputs a signal to the reference direct converter 238 , where the reference signal is converted to a digitized, baseband signal.
- the reference direct converter 238 creates a digitized reference of the spectral energy of the transmitter 280 within the bandwidth 402 of the receiver (FIG. 4) to produce an output, Z(n).
- the reference direct converter 238 transfer the signal Z(n) to the adaptive digital transversal filter 234 and the cross correlation measurement module 240 .
- the adaptive digital transversal filter 234 compensates for the amplitude and phase differences between the transmitter-to-receiver leakage path (the receiver filter 214 and direct converter 220 ) and the reference transmit signal path (the reference filter 236 and direct converter 238 ) used as a reference over the receiver bandwidth or bandwidth of interest.
- the adaptive digital transversal filter 234 aligns and shapes the digitized transmit signal reference, Z(n), from the reference path in amplitude and phase to substantially match the amplitude and phase of the transmit leakage signal through the leakage receiver path.
- the adaptive digital transversal filter 234 outputs an output signal, U(n), to the combiner 232 .
- the combiner 232 uses two summers 424 , 426 to coherently subtract the output, U(n), of the adaptive digital transversal filter 234 from the digitized receiver output, X(n), to form a signal, Y(n), whose transmitter spectral spillage signal power (the error) within the receiver bandwidth is suppressed, as shown in FIG. 4.
- the signal Y(n) is passed to the matched or channel filter 250 , which outputs a signal R(n).
- the signals Y(n) and/or R(n) may be referred to as the ‘residue.’
- the matched or channel filter 250 passes the residue signal R(n) to the cross-correlation measurement module 240 .
- the cross-correlation measurement module 240 identifies the common signal characteristics of the receiver path and the reference path by cross-correlating the signal inputs. In one embodiment, the cross correlation module 240 identifies the common signal characteristics between R(n) and Z(n).
- the cross correlation measurement module 240 outputs parameters C(j) to the canceller adaptation module 910 .
- the canceller adaptation module 910 executes an adaptive cancellation algorithm which is described below and outputs adaptive filter coefficients W(j) to the adaptive digital transversal filter 234 .
- the canceller adaptation module 910 adjusts the adaptive filter coefficients, W(j) such that the cross-correlation between the digitized transmit reference signal, Z(n), and the residue, R(n), are minimized in a least mean-square sense over a bandwidth of interest.
- Y(n) may be used instead of R(n) as the residue, which is input into the cross correlation module 240 .
- the digital adaptive coherent spectral canceller 230 is preferably implemented with two high dynamic range direct converters 220 , 238 .
- the direct converters 220 , 238 provide high dynamic range, substantially identical performance and no automatic gain control. Digitization allows use of high performance, easily repeatable digital filters, such as the adaptive digital transversal filter 234 . In general, digital filters are more repeatable than analog filters. The digital filters are advantageously used in the present invention for more effective cancellation.
- the receiver direct converter 220 preferably converts the coupled signal 217 using the same local oscillator signal, VCO RX , as the reference direct converter 238 , because the canceller 230 focuses on the portion of the filtered transmit signal spectrum that is within the receiver bandwidth, as illustrated in FIGS. 2A, 2B and 4 .
- FIG. 5B illustrates an alternate embodiment of a transceiver 200 ′ of FIG. 3 with separate receive and transmit antenna radiators 120 , 122 and an adaptive transmit canceller 230 .
- the transceiver 200 ′ comprises a receiver bandpass filter 214 , a low noise amplifier (LNA) 216 , a receiver direct converter 220 , a digital adaptive coherent spectral canceller 230 , a matched or channel filter 250 , a directional coupler 235 , a transmitter bandpass filter 218 and a transmitter 280 .
- LNA low noise amplifier
- the transmitter 280 receives data and control signals from the digital processor interface.
- the transmitter 280 is coupled to the transmitter bandpass filter 218 , which is coupled to the directional coupler 235 .
- the directional coupler 235 is coupled to the transmitter antenna radiator 122 and the spectral canceller 230 .
- the receiver antenna radiator 120 is coupled to the receiver bandpass filter 214 , which is coupled to the LNA 216 .
- the LNA 216 is coupled to the receiver direct converter 220 , which is coupled to the digital processor interface and the spectral canceller 230 .
- the spectral canceller 230 is coupled to the processor interface and the matched or channel filter 250 , which is also coupled to the digital processor interface.
- the transceiver 200 ′ of FIG. 5B may still experience transmit spectral spillage or leakage into the receive path as the receiving antenna radiator 120 receives the signal energy radiated by the transmitting antenna radiator 122 .
- the transceiver 200 ′ of FIG. 5B can also be configured to cancel interferences from other transmit antennas located in the proximity of the receiver.
- FIG. 6 illustrates one embodiment of a digital coherent spectral canceller 230 within the transceivers 200 , 200 ′ of FIGS. 5A and 5B.
- the canceller 230 comprises a combiner 232 , an adaptive digital transversal filter 234 , a reference bandpass filter 236 , a reference direct converter 238 and a cross correlation measurement module 240 .
- the reference bandpass filter 236 receives a signal from the directional coupler 235 (FIG. 5A) and outputs a signal to the reference direct converter 238 .
- the reference converter 238 is coupled to the adaptive digital transversal filter 234 , the cross correlation measurement module 240 and the digital processor interface (FIG. 5A).
- the adaptive digital transversal filter 234 is coupled to the canceller adaptation module 910 and the combiner 232 , which is coupled to the matched or channel filter 250 (FIG. 5A) and the receiver direct converter 220 (FIG. 5A).
- the combiner 232 may also be coupled to the cross correlation measurement module 240 .
- the cross correlation measurement module 240 is coupled to the matched filter 250 (FIG. 5A) and the canceller adaptation module 910 .
- the reference direct converter 238 creates a digitized reference of the spectral energy of the transmitter 280 within the bandwidth of the receiver to produce an output, Z(n).
- the output of the reference direct converter 238 , Z(n) comprises an in-phase component, Z I (n), and a quadrature component, Z Q (n), as shown in FIG. 6.
- the reference direct converter 238 outputs both the I and Q components of the signal Z(n) to the adaptive digital transversal filter 234 and to the cross correlation measurement module 240 .
- the cross correlation measurement module 240 also receives an I and a Q component of a residue signal R(n) from the matched or channel filter 250 .
- the adaptive digital transversal filter 234 outputs an in-phase component, U I (n), and a quadrature component, U Q (n) (an I and a Q component) of an output signal, U(n), to the combiner 232 .
- the combiner 232 preferably uses a digital delay ⁇ T S 420 , 422 to compensate for differences in time delays between the reference path and the leakage receiver path.
- the digital delay reduces time de-correlation effects and instabilities due to different path lengths and processing times between the receiver and reference paths.
- T S Integer number of sample periods, T S , for delay added to approximately compensate for the difference in delay between the reference and signal paths.
- the adaptation module 910 adjusts the coefficient vector, W(j), of the adaptive digital transversal filter 234 to minimize the transmitter energy within the bandwidth determined by either the bandwidth of the direct converters 220 and 238 or the bandpass filters 214 and 236 .
- ⁇ ⁇ ( ⁇ ) ⁇ ⁇ ⁇ ⁇ 1 ⁇ ⁇ ⁇ 2 ⁇ ⁇ [ X _ ⁇ ( ⁇ ) ⁇ ⁇ - j ⁇ ⁇ ⁇ ⁇ ⁇ ⁇ ⁇ Ts - Z _ ⁇ ( ⁇ ) ⁇ A _ ⁇ ( ⁇ ) ] 2 ⁇ ⁇ F _ ⁇ ( ⁇ ) ⁇ ⁇ 2 ⁇ ⁇ ⁇ ⁇ ( 5 )
- T Sampling period of output of receiver direct converter 220 and reference direct converter 238 (reciprocal of sampling frequency).
- W k-I ( j +1) ⁇ W k-I ( j )+ ⁇ C k-I ( j ) (6)
- W k-Q ( j +1) ⁇ W k-Q ( j )+ ⁇ C k-Q ( j ) (7)
- FIGS. 9A, 9B, 10 A and 10 B are used to illustrate examples of the cancellation obtainable as a function of the complex coefficients (taps) of the adaptive digital transversal filter 234 , where the bandwidth of F ( ⁇ ) is one quarter of the sampling frequency.
- the horizontal axis represents frequency ⁇ 1000/sample rate.
- the vertical axis represents amplitude in decibels (dB).
- FIG. 9A illustrates exemplary effective baseband responses (transfer functions) 430 , 432 or one embodiment of the receive filter 214 and the transmit reference filter 236 of FIG. 5A, where the receiver and reference filters 214 , 236 , H RX (( ⁇ ) and H REF ( ⁇ ), are Chebyshev bandpass filters designed for equal stop and pass bandwidths with different pass band ripples, such as 0.01 dB and 1.0 dB respectively.
- the baseband responses 430 , 432 shown in FIG. 9A may be switched. In other words, either baseband response 430 , 432 shown in FIG. 9A may be associated with the receive filter 214 and/or the transmit reference filter 236 .
- FIG. 9B illustrates 10 log 10
- Curves 434 - 440 represent a different number of taps for the digital coherent spectral 230 of FIG. 6, where the receiver and reference filters 214 , 236 , H RX ( ⁇ ) and H REF ( ⁇ ), are Chebyshev bandpass filters with baseband responses illustrated in FIG. 9A.
- FIG. 9B illustrates 10 log 10
- Curves 434 - 440 represent a different number of taps for the digital coherent
- the line 434 represents the amount of cancellation achieved with an adaptive digital transversal filter 234 of one tap, which does not compensate (equalize) for the frequency dependent mismatch characteristics between the two paths. As shown in FIG. 9B, if the adaptive digital transversal filter 234 uses an increased number of taps, the cancellation performance is significantly improved.
- the filtering of the reference transmitter output signal with a reference filter 236 having substantially the same design as the receiver filter 214 (FIG. 9A) presents at least two advantages.
- it may reduce dynamic range requirements by providing high attenuation to signals outside the receiver band so that the reference direct converter 238 (FIG. 5A) only digitizes the signal within the spectral sidelobes of the transmit signal that fall within the receiver bandwidth.
- This attenuation may significantly reduce the resolution and linearity requirements of the reference direct converter 238 because the sidelobes within the receiver filter's bandwidth will normally be suppressed more than 40 dB below the peak of the transmitter signal.
- the adaptive transversal equalizer 234 may advantageously compensate for any small differences in the transfer functions using a smaller number of taps than required if H RX ( ⁇ ) and H REF ( ⁇ ) in FIG. 5A were not substantially similar.
- FIG. 10A illustrates exemplary cancellation ratio responses 442 , 444 of another embodiment of the receiver filter 214 and the transmit reference filter 236 of FIG. 5A, where the receiver and reference filters, H RX ( ⁇ ) and H REF ( ⁇ ), are Chebyshev bandpass filters designed for different bandwidths and with different pass band ripples, such as 0.5 dB and 1.5 dB respectively.
- the transfer functions 442 , 440 , H RX ( ⁇ ) and H REF ( ⁇ ), of FIG. 10A are less similar than the transfer functions 430 , 432 of FIG. 9A.
- the baseband responses 442 , 444 shown in FIG. 10A may be switched. In other words, either baseband response 442 , 444 shown in FIG. 10A may be associated with the receive filter 214 and/or the transmit reference filter 236 .
- FIG. 10B illustrates 10 log 10
- Curves 446 - 452 represent a different number of taps for the digital spectral canceller 230 of FIG. 6, where the receiver and reference filters 214 , 236 , H RX ( ⁇ ) and H REF ( ⁇ ), are Chebyshev bandpass filters with baseband responses illustrated in FIG. 10A.
- the transfer functions 442 , 440 , H RX ( ⁇ ) and H REF ( ⁇ ) are less similar than the transfer functions 430 , 432 of FIG. 9A, more complexity (taps) may be needed for the adaptive digital transversal filter 234 to achieve substantially the same level of cancellation performance as the performance shown in FIG. 9B.
- FIG. 7 illustrates one embodiment of a direct converter 220 within the transceiver 200 , 200 ′ of FIGS. 5A or 5 B.
- the structure of the reference direct converter 238 is substantially similar to the structure of the receiver direct converter 220 .
- the direct converter 220 comprises an LNA 202 , a divider 205 , a translating delta-sigma modulator 213 for the in-phase path, a translating delta-sigma modulator 211 for the quadrature path, a receiver frequency synthesizer and clock generator 500 , and I & Q decimation filters 300 .
- the LNA 202 and the LNA 216 of FIG. 5A are two separate components.
- the LNA 202 and the LNA 216 represent a single LNA component.
- the divider 205 may be implemented with a variety of active elements and/or passive elements.
- the divider 205 may be implemented as a splitter.
- the output of the LNA 202 can be coupled directly to both of the translating delta-sigma modulators 211 , 213 .
- the LNA 202 of the direct converter 220 and/or the LNA 216 receives an RF signal 201 from the receiver bandpass filter 214 (FIG. 5A).
- the LNA 202 is coupled to the divider 205 , which is coupled to both translating delta-sigma modulators 213 , 211 .
- the translating delta-sigma modulators 213 , 211 are coupled to the I & Q decimation filters 300 and the receiver synthesizer and clock generator 500 .
- the receiver synthesizer and clock generator 500 is coupled to the reference direct converter 238 (FIG. 5A) and the digital processor interface (FIG. 5A).
- the decimation filters 300 are coupled to the combiner 232 (FIG. 5A).
- the LNA 202 amplifies the incoming signal 201 while avoiding the addition of excessive noise and distortion.
- the divider 205 separates the amplified signal into in-phase and the quadrature components and outputs the in-phase and quadrature components to the in-phase and quadrature modulators 213 , 211 , respectively.
- the modulators 213 , 211 translate the received I & Q signal components by f RX to the baseband of the received signal and digitize the I & Q signal components.
- the modulators 213 , 211 produce outputs to the decimation filter 300 .
- the digital data output of each of the modulators 213 , 211 is a one-bit data stream at the same rate equal to f RX .
- the decimation filter 300 performs digital filtering and decimation to produce digital words at the rate of a clock, CLK.
- the decimation filter 300 converts the stream of 1-bit digital words into a stream of N-bit digital words that comprise conventional binary representations of the signals.
- the decimation filter 300 produces a clock signal, CLK_M, which indicates the rate at which the binary representations are created and is used to transfer the filtered output.
- CLK_M clock signal
- the rate at which the words produced by the decimation filter 300 is determined by the oversampling ratio.
- the output clock rate is 1/M times the rate of the incoming data, or alternatively stated, for the embodiment of FIG. 13, the output clock rate is equal to f RX /M where f RX is the rate of the translation and conversion clock, CLK.
- the decimator filter 300 also attenuates unwanted signals and noise sources outside of the bandwidth of interest while preserving the signals of interest.
- the desired filter characteristics of the decimation filter 300 are selected on the basis of the characteristics of the receive signal as well as the conversion clock rate, f RX .
- the decimation filter 300 has a programmable characteristic which can be modified to accommodate a variety of different waveforms.
- the decimation filer 300 is preferably implemented with a finite impulse response (FIR) filter whose characteristics are modified by changing the value of the filter coefficients. The value of the filter coefficients of a digital filter can be readily changed via software making such modifications practical.
- FIR finite impulse response
- FIG. 8 illustrates another embodiment of a direct converter 800 within the transceiver 200 , 200 ′ of FIGS. 5A or 5 B.
- the direct converter 800 comprises an LNA 802 , a pair of amps 808 , 809 , a pair of balanced mixers 810 , 811 , a pair of low pass filter/amps 812 , 813 , a pair of analog-digital converters 814 , 815 , a quadrature hybrid module 805 and a receiver frequency synthesizer and VCO 816 .
- the LNA 802 and the LNA 216 of FIG. 5A are two separate components
- the LNA 802 and the LNA 216 represent a single LNA component.
- the LNA 802 and/or the LNA 216 receives the RF signal 201 from the receiver bandpass filter 214 (FIG. 5A).
- the LNA 202 is coupled to the divider 804 , which is coupled to both amps 808 , 809 .
- the amp 808 is coupled to the balanced mixer 810 , which is coupled to the quadrature hybrid module 805 and the low pass filter/amp 812 .
- the low pass filter/amp 812 is coupled to the analog-digital converter 814 , which is coupled to the analog digital convert 815 , the combiner 232 (FIG. 5A) and the digital processor interface (FIG. 5A).
- the amp 809 is coupled to the balanced mixer 811 , which is coupled to the quadrature hybrid module 805 and the low pass filter/amp 813 .
- the low pass filter/amp 813 is coupled to the analog-digital converter 815 , which is coupled to the analog digital converter 814 , the combiner 232 (FIG. 5A) and the digital processor interface (FIG. 5A).
- FIG. 11 illustrates another embodiment of a direct converter 220 ′ within the transceiver 200 , 200 ′ of FIGS. 5A or 5 B.
- the direct converter 220 ′ comprises a divider 205 , a translating delta-sigma modulator 213 for the in-phase path, a translating delta-sigma modulator 211 for the quadrature path, a clock generator 504 , an I & Q decimation filters 300 and an I & Q gain quadrature and offset correction module 410 .
- the direct converter 220 ′ of FIG. 11 may include an input LNA coupled to the RF input of the divider 205 .
- the divider 205 receives an RF input from the LNA 216 (FIG. 5A).
- the divider 205 is coupled to both translating delta-sigma modulators 213 , 211 .
- the translating delta-sigma modulators 213 , 211 are coupled to the I & Q decimation filters 300 and the clock generator 504 .
- the clock generator 504 is coupled to the reference direct converter 238 (FIG. 5A) and the digital processor interface (FIG. 5A).
- the decimation filters 300 are coupled to the I & Q gain quadrature and offset correction module 410 , which is coupled to the combiner 232 (FIG. 5A).
- FIG. 12 illustrates another embodiment of a direct converter 800 ′ within the transceiver 200 , 200 ′ of FIGS. 5A or 5 B.
- the direct converter 800 ′ comprises a divider 804 , a pair of amps 808 , 809 , a pair of balanced mixers 810 , 811 , a pair of low pass filter/amps 812 , 813 , a pair of analog-digital converters 814 , 815 , a quadrature hybrid module 805 and an I & Q gain, quadrature and offset correction module 850 .
- the direct converter 800 ′ of FIG. 12 may include an input LNA coupled to the RF input of the divider 205 .
- the divider 804 receives an RF input from the LNA 216 (FIG. 5A).
- the divider 804 is coupled to both amps 808 , 809 .
- the amp 808 is coupled to the balanced mixer 810 , which is coupled to the quadrature hybrid module 805 and the low pass filter/amp 812 .
- the loss pass filter/amp 812 is coupled to the analog-digital converter 814 , which is coupled to the analog digital converter 815 , the I & Q gain, quadrature and offset correction module 850 and the digital processor interface (FIG. 5A).
- the amp 809 is coupled to the balanced mixer 811 , which is coupled to the quadrature hybrid module 805 and the low pass filter/amp 813 .
- the low pass filter/amp 813 is coupled to the analog-digital converter 815 , which is coupled to the analog digital converter 814 , the I & Q gain, quadrature and offset correction module 850 and the digital processor interface (FIG. 5A).
- FIG. 13 illustrates one embodiment of a translating delta-sigma modulator 211 within the direct converters 220 , 220 ′ of FIGS. 7 and 11. Because the modulator 213 is substantially similar to the modulator 211 , only the modulator 211 will be described herein.
- the modulator 211 , 213 comprises a complementary amplifier 310 , a switch 312 , a loop amplifier 314 , a one-bit digital-to-analog converter 316 , a loop filter 318 and an edge-triggered comparator 320 .
- the output 206 of the divider 205 (FIGS. 7 or 11 ) is input into the complementary amplifier 310 , which is coupled to the switch 312 .
- the switch 312 is coupled to the loop amplifier 314 .
- the loop amplifier 314 is coupled to the digital-to-analog converter 316 and the loop filter 318 , which is coupled to the comparator 320 .
- the CLK signal is coupled to the switch 312 and the edge-triggered comparator 320 .
- the modulator 211 translates the received I & Q signal components by an amount equal to the frequency of the CLK signal, f RX , to the baseband of the received signal and digitizes the I & Q signal components.
- the complementary amplifier 310 receives the modulated RF carrier signal. At a non-inverting output, the complementary amplifier 310 produces a voltage that is G times the voltage at the input to the complementary amplifier 310 . At an inverting output, the complementary amplifier 310 produces a voltage that is ⁇ G times the voltage at the input to the complementary amplifier 310 .
- the inverting and non-inverting outputs of the complementary amplifier 310 are coupled to two input ports of a switch 312 .
- the control port of the switch 312 determines which input port of the switch 312 is coupled to the output port of the amplifier 310 and is driven by the conversion clock, CLK, such that the output port of the switch 312 is alternately coupled to the inverting and non-inverting outputs of the complementary amplifier 310 .
- the complementary amplifier 310 and the switch 312 perform the functions of a commutator which inverts the polarity of the modulated RF carrier signal on every half cycle of the conversion clock, CLK. If the frequency of the conversion clock CLK is chosen to be approximately equal to the carrier frequency of the modulated RF carrier signal, effectively, the commutator translates the modulation of the carrier signal down to D.C. centered or frequency offset baseband. In addition to the low frequency signal components, high frequency signal components are also generated by the commutator. However, the high frequency components are attenuated by the delta-sigma modulator 211 and further filtering. In one embodiment, the frequency of the conversion clock CLK is programmable to permit the translation of a variety of waveforms over a range of center frequencies.
- the commutator comprised of the complementary amplifier 310 and the switch 312 is not a conventional downconverter.
- the mathematical paradigm for a conventional downconverter is multiplication by a sinusoidal signal.
- Practical implementations of conventional downconverters (such as circuits employing using diode rings or Gilbert multiplier circuits) are incapable of realizing this mathematical paradigm without introduction of distortion and feed-through effects that result in the creation of undesired spurious signals.
- the mathematical paradigm of the commutator is that of alternately multiplying the input signal by +1 or ⁇ 1 on opposite half cycles of a clock signal.
- Practical implementations of the commutator employing a fast switch behave more closely to this mathematical paradigm, thus avoiding the production of non-linear components of the signal in the baseband signal in comparison to a conventional down converter.
- the output of the switch 312 is coupled to the input of the core delta-sigma modulator, which comprises a loop amplifier 314 , a loop filter 318 , an edge-triggered comparator 320 and a one-bit digital-to-analog (D/A) converter 316 .
- the core delta-sigma modulator is operated at the same frequency as the commutator. Use of a conversion clock operating at or near the carrier frequency provides a significant oversampling ratio in typical embodiments and, hence, leads to high resolution, high dynamic range performance according to well-known principles of delta-sigma conversion.
- the output of the loop amplifier 314 is the difference between the voltage coupled to its non-inverting input port and its inverting input port times a voltage gain, A V , where the voltage gain is typically a large positive constant.
- the loop filter 318 is typically an analog low pass filter but can be embodied in other forms. In one embodiment, the loop amplifier 314 and loop filter 318 act as an integrator. When the voltage value at the signal input to the edge-triggered comparator 320 is greater than a predetermined threshold value at the time the conversion clock transitions, the output is a logic value 1 . When the voltage value at the signal input to the edge-triggered comparator 320 is less than the predetermined threshold value at the time the conversion clock transitions, the output is a logic value 0.
- the output of the edge-triggered comparator 320 is coupled to the input of the one-bit digital-to-analog converter 316 .
- the one-bit digital-to-analog converter 316 produces one of two analog levels at its output depending upon the digital logic value applied to its input.
- the output of the one-bit digital-to-analog converter 316 is coupled to the inverting input of the loop amplified 314 .
- the core delta-sigma modulator 211 , 213 shown in FIG. 13 and described above is a standard, one-bit digital-sigma modulator.
- a variety of delta-sigma modulators and delta-sigma modulation techniques can be combined with the teachings of the present invention.
- additional information concerning delta-sigma modulators is found in Delta - Sigma Data Converters: Theory, Design, and Simulation by Steven R. Norsworthy, published by IEEE Press in 1996.
- MOS metal-oxide semiconductor
- prior art systems use switched-capacitor technology to implement delta-sigma modulators.
- switched-capacitor filters cause aliasing and hence, additional interference to the system.
- MOS switched-capacitor circuits must be operated at a much lower oversampling ratio, they do not have as much resolution for any given order of the delta-sigma modulator, compared to one embodiment of the modulator 211 of the present invention.
- prior art systems typically use higher order loop filters which are only conditionally stable. As the order of the delta-sigma modulator is increased, the implementation of a stable loop that is capable of operating at high clock frequencies becomes more difficult.
- one embodiment of the modulator 211 comprises a continuous-time filter for the loop filter 318 .
- one embodiment of the delta-sigma modulator 211 operates at or near the carrier frequency. Due to the use of a high frequency clock, the use of higher order filtering is not needed to achieve a high degree of resolution. Therefore, the use of a lower order, continuous-time filter is practical in conjunction with one embodiment of the modulators 211 , 213 .
- Continuous-time filters are less difficult and bulky to implement than switched-capacitor filters.
- continuous-time filters can be operated at much higher frequencies than switched-capacitor circuits in a given semiconductor technology.
- the use of a continuous-time filter has the added advantage of eliminating aliasing that is potentially produced by switched-capacitor filters.
- modem delta-sigma converters are currently available that are implemented in silicon metal oxide semiconductor (MOS) technology. Typically such designs use switched capacitor techniques to sample the incoming signal for conversion.
- MOS silicon metal oxide semiconductor
- circuits capable of processing high frequency input signals such as those formed from silicon bipolar, silicon germanium (SiGe), or gallium arsenide (GaAs) technologies, can use current steering architectures in order to increase system efficiencies.
- FIGS. 14 A- 14 E are spectral plots used to illustrate the operation of various embodiments of the transceiver 200 of FIG. 1.
- An understanding of the desired characteristics of the decimation filter 300 can be understood with reference to FIG. 14A where the vertical axis represents energy such as in units of decibels and the horizontal axis represents frequency such as in units of Gigahertz.
- FIG. 14A is a spectral plot showing received signal energies 330 , 332 and 334 centered about three different carrier frequencies, F c1 , F c2 and f c3 , respectively.
- signal energies 330 , 332 and 334 each comprise an incoming waveform comprising a digitally modulated RF signal of interest.
- the conversion clock, CLK operates at a frequency f RX which is between frequencies f c1 and f c2 .
- the horizontal axis has been segmented so that more of the signal energy can be shown.
- FIG. 14B represents the corresponding output of the switch 313 (excluding noise) of FIG. 13, when the spectrum shown in FIG. 14 is applied thereto.
- the frequency f RX is equal to 1851.4 MHz and the frequencies f c1 , f c2 and f c3 are 1851, 1851.6 and 1852.2 MHz, respectively.
- Each of the signal energies 330 , 332 and 334 have a bandwidth of approximately 100 kHz.
- the signal energies 336 , 338 and 340 correspond to signal energies 330 , 332 and 334 , respectively, and are centered about ⁇ 400 kHz, 200 kHz and 800 kHz, respectively. Note that the signal energy 336 has been translated to the negative portion of the frequency axis.
- a dashed line 342 of FIG. 14C represents the transfer curve of the decimation filter 300 in one embodiment.
- the low pass decimation filter 300 passes all three signal energies 336 , 338 and 340 .
- each of the signal energies 336 , 338 and 340 could be produced by a different transmitting unit.
- none of the signal energies are centered about D.C.
- the effects of any DC offset in the system and 1/f noise denoted by the increase in the spectral noise density curve 343 around zero frequency
- the decimation filter 300 is implemented with low pass filtering and equivalent bandpass filtering is implemented in the following matched filter.
- the decimation filter 300 is more frequency selective such that only one of the signal energies (such as might be produced by a single transmitting unit) is passed without substantial attenuation.
- the dashed line 344 shows such a decimation filter 300 transfer characteristic.
- only the signal energy 338 is efficiently passed through the decimation filter 300 .
- the down-converted waveform is centered about the D.C., i.e., the waveform has zero frequency offset, as shown in FIG. 14E.
- Conversion to D.C. centered baseband has the benefit of achieving higher resolution for a given clock rate which can be a particular benefit for wide band signals where the effects of quantization noise should be minimized.
- the effects of 1/f noise are less pronounced in a wide band system and can be filtered with a notch filter at zero frequency without significantly degrading the performance.
- a dashed line 346 of FIG. 14E represents the transfer curve of the decimation filter 300 in one such embodiment. More information concerning the design of decimation filters can be found in Multi - Rate Digital Signal Processing , Prentice-Hall, Inc., Englewood Cliffs, N.J., 1983 by R. E. Crochiere and L. R. Rabiner.
- the receiver 106 (FIG. 1) to operate in accordance with more than one communication protocol.
- the receiver can operate in a narrow band time division multiple access (TDMA) system such as Global System for Mobile Communication (GSM) or a wide band code division multiple access (CDMA) system such as defined in the Telephone Industry Association, Electronic Industry Association (TIA/EIA) interim standard entitled “Mobile Station—Base Station Capability Standard for Dual-Mode Wide band Spread Spectrum Cellular System,” TIA/EIA/IS-95.
- TDMA time division multiple access
- GSM Global System for Mobile Communication
- CDMA wide band code division multiple access
- the decimation filter 300 can take on a narrow band transfer characteristic, shown by dashed line 344 .
- the offset and bandwidth of this filter shown by line 342 , may be increased according to well known principles of digital filtering and signal reception.
- a single wide band, low-pass decimation filter could be utilized and the programmable bandwidth implemented in the following matched filtering.
- FIG. 11 the output of the decimation filter 300 is the input into the I & Q gain quadrature and offset correction component 410 .
- FIG. 15 is a block diagram showing one embodiment of the component 410 .
- the clock and data output of the decimation filters 300 are coupled to a calibration circuit 350 .
- the calibration circuit 350 adjusts the relative gain and phase so that the in-phase and quadrature signal paths are balanced with respect to each other. In order to avoid introduction of distortion into the signals, it is important that the relative gain and phase of the in-phase and quadrature signal paths are the same.
- One advantage of a digital signal processing architecture shown in FIG. 15 is that the parameters can be controlled in the digital circuit elements more easily than in analog circuit elements. Typically, unbalances originate from the differences in gain between the I and Q channels and errors in the relative 90° phase shift between the I and Q channels. Additionally, any differences in the DC offsets can be calibrated out. Additional information concerning accomplishment of calibration can be found in U.S. Pat. No. 5,422,889 entitled “OFFSET CORRECTION CIRCUIT,” and in U.S. Pat. No.
- the output of the calibration circuit 350 is coupled to the input of a sampling rate converter 352 .
- the sampling rate converter 352 converts and synchronizes the data rate of the signal to the rate of an external clock, CLK waveform .
- this function is accomplished with a linear or higher order interpolation method such as the one described in “Advanced Digital Signal Processing” by J. G. Proakis, et al., and McMillan Publishing Co.
- the output of the sampling rate converter 352 is coupled to the input of a frequency translator 354 .
- the frequency translator 354 is used to translate the center frequency of the signal of interest to a D.C. centered baseband.
- the frequency translator 354 multiplies the signal at the output of the sampling rate converter 352 with a digital representation of a sinusoidal signal having a frequency equal to the center of the frequency of signal of interest.
- the advantage of frequency translation is that it allows the matched filter 356 for the signal to be implemented as a low pass filter and provides the baseband I and Q inputs required for the digital demodulator input. For the situation shown in FIG. 14E where there is only one signal of interest and it has zero offset, the frequency translator 354 is preferably not used.
- the frequency translator 354 When the frequency translator 354 is utilized, and the output of the matched or channel filter 250 is utilized as an input to the cross correlation measurement module 240 as shown in FIG. 5A, then the input Z(n) to the cross correlation measurement module 240 must also undergo the same frequency translation.
- the translation can be accomplished by including a frequency translator in the cross correlation measurement module 240 .
- the output of the frequency translator 354 is coupled to a low pass filter 356 which can operate as a signal matched filter.
- the low pass filter 356 is also used to reject interference outside the bandwidth of interest.
- the output of the low pass filter 356 provides a digital I and Q signal input to the digital demodulator that is synchronized with the digital demodulator clock—CLK WAVEFORM .
- FIG. 16 is a block diagram showing one embodiment of the clock generator 500 of FIG. 7.
- a frequency synthesizer 360 produces an analog waveform at twice the rate of the conversion clock, CLK.
- the output of the frequency synthesizer 360 is coupled to the input of a limiting amplifier 362 .
- the positive going zero crossing of the signal output by the frequency synthesizer 360 is compared to a threshold by the limiting amplifier 362 .
- the threshold is chosen appropriately, the limiting amplifier 362 produces a waveform with digital logic values at the same frequency as that of the output from the frequency synthesizer 360 and having a 50% duty cycle (i.e., the duration of the logic “1” pulse is the same as the duration of the logic “0” pulse).
- the limiting amplifier 362 drives a master slave flip-flop 376 comprising a master latch 364 and a slave latch 368 .
- the master-slave flip-flop 376 is configured in a divide-by-two configuration. In this configuration, a Q output 366 and a ⁇ overscore (Q) ⁇ output 372 of the flip-flop 364 are connected to the D and ⁇ overscore (D) ⁇ inputs of the flip-flop 368 , respectively, and a Q output 370 and a ⁇ overscore (Q) ⁇ output 374 of the flip-flop 368 are connected to the ⁇ overscore (D) ⁇ and D inputs of the flip-flop 364 , respectively.
- the four latch outputs 366 , 372 , 370 and 374 have clock phases of 0°, 90°, 380°, and 270°, with respect to one another.
- Two of these outputs (for example, output 366 and output 370 ) can be used as I_CLK and Q_CLK, respectively.
- FIG. 16 is included explicitly herein for illustration purposes, a variety of other means (such as a ring oscillator) can be used to generate a clock signal in accordance with the present invention.
- FIG. 17 illustrates another embodiment of a translating delta-sigma modulator 380 which employs double-sampling (i.e., samples on both edges of a clock signal).
- the delta-sigma modulator 380 of FIG. 17 operates under some of the same principles as the single sampled architecture shown in FIG. 13 while doubling the sample rate, thereby relaxing the speed requirements for circuitry by a factor of two.
- the delta-sigma modulator 380 can be used as within the architecture shown in FIG. 7 as the transmitting delta-sigma modulators 211 and 213 .
- complementary amplifier 382 receives the digitally modulated RF signal centered about the carrier frequency. At a non-inverting output, the complementary amplifier 382 produces a voltage that is G times the voltage at the input to the complementary amplifier 382 . At an inverting output, the complementary amplifier 382 produces a voltage that is ⁇ G times the voltage at the input to the complementary amplifier 382 .
- the inverting and non-inverting outputs of the complementary amplifier 382 are coupled to two input ports of a switch 384 .
- the control port of the switch 384 determines which input port is coupled to the output port and is driven by the conversion clock, CLK, such that the output port of the switch 384 is alternately coupled to the inverting and non-inverting outputs of the complementary amplifier 382 .
- the complementary amplifier 382 and the switch 384 perform the functions of a commutator as explained more fully above with reference to FIG. 13.
- the output of the switch 384 is coupled to the input of the core double-sampling delta-sigma modulator.
- the core double-sampling delta-sigma modulator is comprised of a combiner 388 , a loop amplifier 390 , a loop filter 392 , an even-phase edge-triggered comparator 394 A, an odd-phase edge-triggered comparator 394 B, an even-phase digital-to-analog converter 396 A and an odd-phase digital-to-analog converter 396 B.
- the output of the switch 384 is coupled to the non-inverting input of the loop amplifier 390 .
- the output of the loop amplifier 390 is the difference between the voltage coupled to its non-inverting input port and its inverting input port times a voltage gain A v where the voltage gain is typically a large positive constant.
- the output of the loop amplifier 390 is coupled to the input of the loop filter 392 .
- the loop filter 392 is an analog low pass filter but can be embodied in other forms.
- the loop amplifier 390 and loop filter 392 act as an integrator.
- the output of the loop filter 392 is coupled to the input of the even-phase edge-triggered comparator 394 A and also to the input of the odd-phase edge-triggered comparator 394 B.
- the clock inputs of the even-phase edge-triggered comparator 394 A and the odd-phase edge-triggered comparator 394 B are coupled to the conversion clock, CLK.
- the even-phase edge-triggered comparator 394 A and odd-phase edge-triggered comparator 394 B are clocked using opposite edges of the comparison clock, CLK.
- the even-phase edge-triggered comparator 394 A performs a comparison on the rising edge of the comparison clock, CLK
- the odd-phase edge-triggered comparator 394 B performs a comparison on the falling edge of the comparison clock, CLK.
- the logic values output by the even-phase edge-triggered comparator 394 A and the odd-phase edge triggered comparator 394 B are coupled to the input of the digital-to-analog converter 396 A and the digital-to-analog converter 396 B, respectively.
- the outputs of the digital-to-analog converter 396 A and digital-to-analog converter 396 B are combined through the combiner 388 and drive the inverting input of the loop amplifier 390 .
- the combiner 388 simply adds the two values together.
- the combiner 388 time-division multiplexes the values into the loop.
- One useful attribute of the first embodiment of the combiner 388 is that linearity can be achieved without tight matching between the tight-to-analog converter 396 A and the digital-to-analog converter 396 B since their respective outputs are effectively averaged before being presented to the loop amplifier.
- the outputs of edge-triggered comparator 394 A and edge-triggered comparator 394 B are also coupled to the decimation filter 300 in a similar manner to the single-sampled case.
- typically the architecture of the decimation filter 300 is appropriately modified to accommodate processing the samples in the form of two bit serial words instead of a single high speed serial bit stream.
- the direct converter embodiments are not limited in dynamic range in contrast to conventional multistage down converters.
- the amplitude of the incoming waveform applied to the receiver direct converter 220 is in fixed proportion of an amplitude of a signal received by the receive antenna radiator 102 because no automatic gain control mechanism is included.
- automatic gain control used to extend the dynamic range of a receiver may be undesirable for spectrally crowded applications such as cellular communications because the automatic gain control may make the receiver sensitivity dependent upon signals and interference that are outside the signal channel.
- the digital decimation filtering of the receiver direct converter 220 (FIGS. 5A, 7 and 8 ), the reference direct converter 238 (FIG. 5A), and the matched or channel filter 250 (FIG. 5A) with low side lobes may significantly suppress the transmit signal outside of the desired signal bandwidth.
- these digital decimation filters 220 , 238 , 250 have approximately 90 dB attenuation. This attenuation advantageously reduces the amount of filtering provided by the duplexer receiver filter 214 .
- the reference direct converter 238 and/or the receiver direct converter 220 have a sampling rate approximately equal to that of the carrier frequency of interest. This may significantly reduce the requirements on the duplexer receiver filter 214 and the reference bandpass filter 236 to provide attenuation at the aliasing frequency (i.e., at half the RF frequency).
Abstract
A method and apparatus for attenuating a transmit signal spectrum in a receiver bandwidth. The apparatus relates to an adaptive, digital, coherent, spectral canceller. The method comprises digitizing both a corrupted receiver signal and a reference transmit signal and then digitally implementing an adaptive coherent spectral canceller module to suppress a residue transmit spectral signal power within the receiver bandwidth.
Description
- This application is a continuation of, and claims priority from U.S. patent application Ser. No. 09/487,396, filed Jan. 18, 2000, which is incorporated in its entirety by reference herein.
- 1. Field of the Invention
- The present invention relates to wireless communications. More specifically, the invention relates to canceling transmitter signal energy in a receiver bandwidth.
- 2. Description of the Related Art
- There is a growing need for low cost, high performance radio transceivers that can operate in a full duplex mode in support of such applications as code division multiple access (CDMA), global system for mobile communications (GSM) and time division multiple access (TDMA) remote units and base stations.
- Currently, a standard transceiver comprises a transmitter, a receiver and a duplexer. One problem with standard transceivers is some of the transmit signal leaks into the receiver, which corrupts the received signal processed by the receiver.
- In some transceivers, the duplexer functions to reduce the transmit signal received by the receiver through the use of one or more circulators, receive filters and/or transmit filters. For example, a circulator within the duplexer directs the transmit signal toward an antenna radiator, while a receive filter rejects the residual transmit energy directed toward the receiver that falls outside of the receive filter bandwidth. Alternatively, some duplexers do not use a circulator.
- Imperfect duplexers create two problems. First, the high-level transmit signal from the transmitter drives the receiver into a non-linear operating region. This problem can be solved by increasing the rejection characteristics of the receive filter to more effectively attenuate the main spectral lobe of the transmitter signal. Alternatively, this problem may be mitigated through the use of a high dynamic range, direct conversion, digital receiver that can implement the requisite filter characteristics with lossless digital filtering.
- A second problem is noise from the transmitter leaks into the receiver and raises its noise figure. The second problem can be mitigated by further filtering the transmitter output and by reducing the spectral sidelobe energy through signal design and linearization of a transmit amplifier. These changes, however, are undesirable from the standpoint of transmitter efficiency.
- Canceling an undesired signal with an adaptive canceller was introduced by B. Widrow et al. inAdaptive Noise Canceling Principles and Applications, Proc. IEEE, Vol. 63, pp. 1269-1716, December 1975, which is hereby incorporated herein by reference. Specific applications for adaptive noise canceling and echo cancellation in data transmission over telephone channels are given by J. G. Proakis et al. in Advanced Digital Signal Processing, Macmillan Publishing Co., New York, 1992, pp. 322-327, 331-332.
- The present invention relates to a method and apparatus for adaptive digital cancellation of a transmit signal spectrum in a receiver bandwidth. The apparatus relates to a digital coherent spectral canceller that attenuates the spectral components from the transmitter that fall within the bandwidth of the receiver. The digital adaptive coherent spectral canceller digitizes both a corrupted receiver signal and a reference transmit signal and then digitally implements an adaptive coherent spectral canceller adaptation module.
- One embodiment of the invention is implemented with a direct conversion digital receiver, which achieves an image-free, high dynamic range without the use of automatic gain control. Automatic gain control used to extend the dynamic range of a receiver may be undesirable for spectrally crowded applications such as cellular communications because the automatic gain control may make the receiver sensitivity dependent upon signals and interference that are outside the signal channel. For example, it is possible for a strong signal in an adjacent channel to capture the receiver front end desensitize the receiver such that a weak signal in the channel of interest is undetectable. This is particularly harmful in a base station receiver where the receiver receives incoming signals from multiple remote units. Furthermore, the use of automatic gain control will likely require the digital coherent canceller to track the gain changes which will introduce errors and noise.
- One embodiment of the invention uses an adaptive transversal filter in an all digital implementation to compensate for the amplitude and phase differences between the transmitter-to-receiver leakage path and the transmit signal path utilized as a reference over the receiver bandwidth or bandwidth of interest.
- An advantage of the invention is reduced complexity and cost, which is achieved through: (1) reduced performance requirements of the duplexer, (2) reduction in the requirement for transmit signal filtering, and (3) reduction in the requirements for high linearity or linearization of the transmit amplifier.
- One aspect of the invention relates to an adaptive, coherent, digital canceller system. The canceller system is configured to attenuate a signal spectrum from a transmitter which falls within a bandwidth of a receiver. The canceller system comprises a reference bandpass filter, a reference direct converter, a cross correlation measurer, an adaptation coherent spectral canceller algorithm module executed, for example, on a microcontroller, an adaptive digital transversal filter, and a combiner.
- The reference direct converter is adapted to output a digitized transmit signal reference of a spectral energy of the transmitter within the bandwidth of the receiver. The adaptive digital transversal filter is adapted to align an amplitude and phase of a digitized transmit signal reference in a reference path with a transmit signal in a leakage receiver path. The adaptive digital transversal filter outputs a compensated or equalized digitized transmit signal reference. The combiner is adapted to coherently subtract the compensated, digitized transmit signal reference from a corrupted, digitized receiver signal to form a residue, having transmitter spectral signal power within the bandwidth of a receiver is suppressed.
- Another aspect of the invention relates to a method of attenuating a transmit signal spectrum in a bandwidth of a receiver. The method comprises digitizing a received signal which is corrupted by components of a transmit signal, creating a digitized reference transmit signal of the transmit signal within the bandwidth of the receiver, aligning the digitized reference transmit signal in amplitude, phase and time delay with the digitized received signal, subtracting the digitized reference transmit signal from the digitized received signal to form a residue, and suppressing a transmitter spectral signal power of the residue within the bandwidth of the receiver.
- FIG. 1 illustrates one embodiment of a transceiver and a processor in accordance with the present invention.
- FIG. 2A illustrates transmitter spectral spillage into a receiver bandwidth with a single channel.
- FIG. 2B illustrates transmitter spectral spillage into a receiver bandwidth with multiple channels.
- FIG. 3 illustrates one embodiment of a communication system with a transceiver in accordance with the present invention.
- FIG. 4 illustrates an exemplary spectral density at an input and output of an adaptive coherent spectral canceller within the transceiver of FIG. 1.
- FIG. 5A illustrates one embodiment of a transceiver with a shared transmit and receive antenna radiator.
- FIG. 5B illustrates one embodiment of a transceiver with separate receive and transmit antenna radiators.
- FIG. 6 illustrates one embodiment of a digital coherent spectral canceller within the transceiver of FIGS. 5A and 5B.
- FIG. 7 illustrates one embodiment of a direct converter within the transceiver of FIGS. 5A and 5B.
- FIG. 8 illustrates another embodiment of a direct converter within the transceiver of FIGS. 5A and 5B.
- FIG. 9A illustrates exemplary baseband responses of one embodiment of a receive filter and a transmit reference filter within the transceiver of FIG. 5A.
- FIG. 9B illustrates cancellation ratios of spectral leakage from the transmitter as a result of a digital coherent spectral canceller of FIG. 5A having varying number of taps with the filter responses of FIG. 9A.
- FIG. 10A illustrates exemplary baseband responses of another embodiment of the receive filter and transmit reference filter within the transceiver of FIG. 5A.
- FIG. 10B illustrates exemplary outputs of one embodiment of a digital spectral cancellers of FIG. 5A having varying numbers of taps with the filter responses of FIG. 10A.
- FIG. 11 illustrates another embodiment of a direct converter within the transceiver of FIGS. 5A and 5B.
- FIG. 12 illustrates another embodiment of a direct converter within the transceiver of FIGS. 5A and 5B.
- FIG. 13 illustrates one embodiment of a translating delta-sigma modulator within the direct converter of FIGS. 7 and 11.
- FIGS.14A-14E are spectral plots used to illustrate the operation of various embodiments of the transceiver of FIG. 1.
- FIG. 15 is a block diagram showing one embodiment of a component within FIGS. 11 and 12.
- FIG. 16 is a block diagram showing one embodiment of a clock generator of FIG. 7.
- FIG. 17 illustrates another embodiment of a modulator within the direct converters of FIGS. 7 and 11.
- FIG. 1 illustrates an exemplifying communication system in which the present invention may be implemented. Alternatively, the present invention may be used in other systems and applications. FIG. 1 illustrates one embodiment of a
transceiver 200 and aprocessor 900 in accordance with the present invention. In one embodiment, thetransceiver 200 is a full duplex transceiver, although the invention can be used in communication systems in which the base station concurrently transmits and receives regardless of whether the communication system provides full duplex operation. - In one embodiment, the
transceiver 200 is for code division multiple access (CDMA) mobile and base station transceivers. In another embodiment, thetransceiver 200 is for a time division multiple access (TDMA) transceiver, such as a transceiver used for GSM and IS-54 base stations. In FIG. 1, thetransceiver 200 comprises transmit and receiveantenna radiator 102, aduplexer 210, areceiver 106, atransmitter 280 and a digital coherentspectral canceller 230. In one embodiment, thereceiver 106 is a direct conversion digital receiver described herein. In another embodiment, thereceiver 106 is a modified direct conversion digital receiver described in U.S. Pat. No. 5,557,642, entitled “Direct Conversion Receiver For Multiple Protocols” by Williams modified to exclude automatic gain control. - In FIG. 1, the
transmitter 280 and the digital coherentspectral canceller 230 are coupled to theprocessor 900. The transmit and receiveantenna radiator 102 is coupled to theduplexer 210, which is coupled to thereceiver 106, the digital coherentspectral canceller 230 and thetransmitter 280. Theprocessor 900 sends data and control signals to thetransmitter 280, which sends data to theduplexer 210 for transmission via theantenna radiator 102. Theduplexer 210 sends a reference transmit signal to thecanceller 230. Theduplexer 210 also receives data from theradiator 102 and sends data to thereceiver 106, which sends the received data to thecanceller 230. Thecanceller 230 and theprocessor 900 use the reference transmit signal from theduplexer 210 to cancel noise from thetransmitter 280 that leaked into the receiver bandwidth. - In some conventional systems, noise from the
transmitter 280 that leaks into thereceiver 106 can be mitigated by further filtering the transmitter output and by reducing the spectral sidelobe energy through signal design and linearization of a transmit amplifier. These changes, however, are undesirable from the standpoint of transmitter efficiency. - The digital adaptive coherent
spectral canceller 230 of the present invention attenuates the spectral components from thetransmitter 280 that fall within the bandwidth of thereceiver 106. Thedigital canceller 230 digitizes both a corrupted receiver signal and a reference transmit signal and then digitally implements an adaptive coherent spectral canceller adaptation module with theprocessor 900. - The
receiver 106 is preferably a direct conversion digital receiver that achieves an image-free, high dynamic range without the use of automatic gain control. Automatic gain control used to extend the dynamic range of a receiver may be undesirable for spectrally crowded applications such as cellular communications because the automatic gain control may make the receiver sensitivity dependent upon signals and interference that are outside the signal channel. - The digital coherent
spectral canceller 230 of the present invention and the methods of using thecanceller 230 provide several advantages over conventional systems. These advantages include (1) reducing performance requirements of theduplexer 210, (2) reducing the requirement for transmit signal filtering, and (3) reducing in the requirements for high linearity or linearization of the transmit amplifier. - FIG. 2A illustrates a single-channel system with a transmitted signal power spectrum (spectral density)400, denoted as {overscore (S)}T(f) and centered about a transmit frequency fT, and a received
communication signal power 404, denoted as {overscore (S)}R(f) and centered about a receive frequency fR. In FIGS. 2A, 2B and 4, the horizontal axis represents frequency (f), such as in units of Gigahertz, and the vertical axis represents signal power amplitude, such as in units of decibels. The horizontal axis has been segmented so that more of the signal energy can be shown. - FIG. 2A also illustrates a
receiver bandwidth 402 from a receiver bandpass filter, such as thereceiver bandpass filter 214 described below with reference to FIG. 5A. Thereceiver bandwidth 402 is designed to pass the receivedsignal 404 but not the main transmitsignal 400. The transmittedsignal power spectrum 400 and thereceiver bandwidth 402 are shown in dashed lines. FIG. 2A also shows transmitspectral spillage 406 in thereceiver bandwidth 402 from a single channel. In other words,noise 406 from the transmitter 280 (FIG. 1) leaks (spills) into thereceiver 106 and raises its noise figure. Thecanceller 230 of the FIG. 1 is designed to cancel this transmitnoise 406 within thereceiver bandwidth 402. - FIG. 2B illustrates a multiple-channel system with a plurality of transmitted
signal powers 404′, 404″, 404′″, areceiver bandwidth 402 and a plurality of receivedcommunication signal powers 404′, 404″, 404′″. FIG. 2B also shows transmitterspectral spillage 406 in thereceiver bandwidth 402 from multiple channels (multiple transmit signals and multiple receive signals). In FIGS. 2A and 2B, the plots represent signals which have passed through an amplifier in thereceiver 106 of FIG. 1, such as thelow noise amplifier 216 shown in FIG. 5A. - FIG. 4 illustrates an exemplary spectral density at an input and output of the adaptive coherent
spectral canceller 230, as shown in FIG. 1 and described in greater detail below with reference to FIGS. 5A, 5B and 6. FIG. 4 is based on the plots within thereceiver bandwidth 402 of FIG. 2B. In FIG. 4, thebandpass transfer function 402 of the receiver bandpass filter 214 (FIG. 5A) passes the three receivedsignal power densities 404′, 404″, 404′″ and the transmit noisespectral density 406. - In FIG. 4, a direct converter filter, such as the
direct converter 220 described below with reference to FIG. 5A, has an even moreselective transfer function 408 than the receiver bandpassfilter transfer function 402. The direct converterfilter transfer function 408 passes only two 404′, 404″ of the three signalspectral densities 404′, 404″, 404′″ and a portion of the transmit noisespectral density 406. In other words, only two receivedspectral densities 404′, 404″ are efficiently passed through thedirect converter 220 and on to the canceller 230 (FIGS. 1 and 5A). As shown in FIG. 4, thecanceller 230 reduces the transmit noisespectral density 406 to a residualspectral density 410 or lower. - FIG. 3 illustrates some of the inputs, outputs, signal flow and functions of the
processor 900 and thetransceiver 200 of FIG. 1. FIG. 3 illustrates anantenna radiator 102, atransceiver 200, aprocessor 900 and a transmitter frequency synthesizer andVCO 600. Thetransceiver 200 comprises an adaptivedigital canceller 230, as shown in FIGS. 1, 5A, 5B and 6, in accordance with the present invention. - In FIG. 3, the
processor 900 comprises a cancelleradaptation algorithm module 910, a digital demodulator module, a frequency control module and a signal waveform clock module. The modules may comprise software, firmware, hardware or any combination thereof or may be implemented with discrete logic or in an application specific integrated circuit (ASIC). In an alternative embodiment, thecanceller adaptation module 910 may be executed by a dedicated element, which is separate from theprocessor 900. In one embodiment, theprocessor 900 comprises a collection of processing elements such as an ARM or a Power Point microprocessor and a dedicated digital modem such as the Qualcomm MSM 3000. Alternatively, in other embodiments, the configurations of the embodiments, the configuration of theprocessor 900 may be different. - In FIG. 3, the
antenna radiator 102 is coupled to thetransceiver 200, which is coupled to theprocessor 900. Theprocessor 900 is coupled to the transmitter frequency synthesizer andVCO 600, which is coupled to thetransceiver 200. Theprocessor 900 may be coupled to other components that transfer data and/or control signals to theprocessor 900 or receive signals from theprocessor 900. - As shown in FIG. 3, the
transceiver 200 transmits and receives signals via theantenna radiator 102. Thetransceiver 200 passes various data to theprocessor 900, such as an I & Q filtered signal and correlation measurement data, C(j), as described below with reference to FIG. 5. In this application, the underlining of a variable, such as C(j), denotes that the variable is a complex variable, i.e., the variable consists of an in-phase (I) component and a quadrature component (Q). - The
processor 900 may pass various data to thetransceiver 200, such as a clock waveform, CLK, canceller parameters, W(j), and transmitter data. Theprocessor 900 may also provide control information, such as a receiver frequency control signal, to thetransceiver 200. Theprocessor 900 may also provide control information, such as a transmitter or calibration frequency control signal, to the transmitter frequency synthesizer andVCO 600, which transmits a signal to thetransceiver 200. - FIG. 5A illustrates one embodiment of the
transceiver 200 of FIG. 3 with a shared receiver/transmitter (Rx/Tx)antenna radiator 102 and an adaptive, digital, coherentspectral canceller 230. In FIG. 5A, thetransceiver 200 comprises aduplexer 210, a low noise amplifier (LNA) 216, a receiverdirect converter 220, a digital adaptive coherentspectral canceller 230, a matched orchannel filter 250 and atransmitter 280. Theduplexer 210 comprises acirculator 212, areceiver bandpass filter 214, atransmitter bandpass filter 218 and adirectional coupler 235. Thecanceller 230 comprises acombiner 232, an adaptive digital transversal filter (or equalizer) 234, areference bandpass filter 236, a referencedirect converter 238 and a crosscorrelation measurement module 240. - In another embodiment, the
duplexer 210 does not comprise adirectional coupler 235, and instead some of the energy passed to theantenna radiator 102 is coupled to an input of thereference bandpass filter 236 within thecanceller 230. - In FIG. 5A, the
transmitter 280 receives data and control signals from a digital processor interface. Thetransmitter 280 is coupled to thetransmitter bandpass filter 218, which is coupled to thedirectional coupler 235. Thedirectional coupler 235 is coupled to thecirculator 212 and thereference bandpass filter 236. Thecirculator 212 is coupled to theantenna radiator 102 and thereceiver bandpass filter 214, which is coupled to theLNA 216. TheLNA 216 is coupled to the receiverdirect converter 220, which is coupled to thecombiner 232, the referencedirect converter 238 and the digital processor interface. Thecombiner 232 is coupled to the matched orchannel filter 250 and the adaptive digitaltransversal filter 234. In one embodiment, thecombiner 232 is also coupled to the crosscorrelation measurement module 240. The matchedfilter 250 is coupled to the digital processor interface and the crosscorrelation measurement module 240. Thereference bandpass filter 236 is coupled to the referencedirect converter 238, which is coupled to the receiverdirect converter 220, the digital processor interface, the adaptive digitaltransversal filter 234 and the crosscorrelation measurement module 240. Both the adaptive digitaltransversal filter 234 and the crosscorrelation measurement module 240 are coupled to thecanceller adaptation module 910, which is executed by the digital processor 900 (FIG. 3) or by a separate microcontroller. - The general use and operation of the
transceiver 200 is described with reference to FIGS. 1-5A. In FIG. 5A, theantenna radiator 102 receives an incoming signal and transfers the incoming signal to thecirculator 212. In one embodiment, the incoming signal is a high frequency signal, such as a digitally modulated RF signal centered about a carrier frequency of approximately 2 Gigahertz (GHz). Alternatively, in other embodiments, other signals and carrier frequencies may be used in accordance with the invention. Thecirculator 212 transfers the received signal to thereceiver bandpass filter 214. - Concurrently, the
transmitter 280 may be transmitting a signal to thetransmitter bandpass filter 218, which outputs a transmitter signal STX(t), to thecirculator 212. The signals described herein may be represented as either functions of time, (t), or alternatively, as their Fourier equivalents, functions of frequency, (ω). Thecirculator 212 directs the transmit signal to theantenna radiator 102. As shown in FIGS. 2A, 2B and 4, some of the energy associated with the transmit signal, STX(t), leaks into thereceiver bandpass filter 214 because thecirculator 212 does not provide perfect isolation. Thisleakage signal energy 406 is limited by thereceiver bandpass filter 214, as shown in FIGS. 2A, 2B and 4. - The resultant output of the
receiver bandpass filter 214 due to STX(t) and the received signal is input to theLNA 216. TheLNA 216 amplifies the signal and passes the signal to the receiverdirect converter 220. The receiverdirect converter 220 converts the filteredsignal 217 to a baseband, digitized and filtered signal to produce an output, X(n). - In FIG. 5A, the digital coherent
spectral canceller 230 adaptively suppresses the transmitterspectral spillage 406 in thereceiver band 402 shown in FIG. 4. Specifically, thedirectional coupler 235 provides a reference signal to the digital coherentspectral canceller 230. The reference signal represents the transmit signal, STX(t). Within thecanceller 230, the reference signal is passed to thereference bandpass filter 236. The transfer function HREF(ω), of thereference bandpass filter 236 preferably matches the transfer function HRX(ω), of thereceiver bandpass filter 214 in the previously described leakage path to the receiverdirect converter 220. Alternatively, one of thefilters other filter reference bandpass filter 236 outputs a signal to the referencedirect converter 238, where the reference signal is converted to a digitized, baseband signal. - The reference
direct converter 238 creates a digitized reference of the spectral energy of thetransmitter 280 within thebandwidth 402 of the receiver (FIG. 4) to produce an output, Z(n). The referencedirect converter 238 transfer the signal Z(n) to the adaptive digitaltransversal filter 234 and the crosscorrelation measurement module 240. - The adaptive digital
transversal filter 234 compensates for the amplitude and phase differences between the transmitter-to-receiver leakage path (thereceiver filter 214 and direct converter 220) and the reference transmit signal path (thereference filter 236 and direct converter 238) used as a reference over the receiver bandwidth or bandwidth of interest. In a preferred embodiment, the adaptive digitaltransversal filter 234 aligns and shapes the digitized transmit signal reference, Z(n), from the reference path in amplitude and phase to substantially match the amplitude and phase of the transmit leakage signal through the leakage receiver path. The adaptive digitaltransversal filter 234 outputs an output signal, U(n), to thecombiner 232. - In FIG. 5A, the
combiner 232 uses twosummers transversal filter 234 from the digitized receiver output, X(n), to form a signal, Y(n), whose transmitter spectral spillage signal power (the error) within the receiver bandwidth is suppressed, as shown in FIG. 4. The signal Y(n) is passed to the matched orchannel filter 250, which outputs a signal R(n). The signals Y(n) and/or R(n) may be referred to as the ‘residue.’ - The matched or
channel filter 250 passes the residue signal R(n) to thecross-correlation measurement module 240. Thecross-correlation measurement module 240 identifies the common signal characteristics of the receiver path and the reference path by cross-correlating the signal inputs. In one embodiment, thecross correlation module 240 identifies the common signal characteristics between R(n) and Z(n). The crosscorrelation measurement module 240 outputs parameters C(j) to thecanceller adaptation module 910. - The
canceller adaptation module 910 executes an adaptive cancellation algorithm which is described below and outputs adaptive filter coefficients W(j) to the adaptive digitaltransversal filter 234. Thecanceller adaptation module 910 adjusts the adaptive filter coefficients, W(j) such that the cross-correlation between the digitized transmit reference signal, Z(n), and the residue, R(n), are minimized in a least mean-square sense over a bandwidth of interest. - Alternatively, in another embodiment, Y(n) may be used instead of R(n) as the residue, which is input into the
cross correlation module 240. The cancellation bandwidth can be expanded to the bandwidth of thereceiver bandpass filter 214 andreference bandpass filter 236, F(ω)=1, by utilizing the output of thecombiner 232, Y(n), for the residual signal instead of R(n). - The digital adaptive coherent
spectral canceller 230 is preferably implemented with two high dynamic rangedirect converters direct converters transversal filter 234. In general, digital filters are more repeatable than analog filters. The digital filters are advantageously used in the present invention for more effective cancellation. - In FIG. 5A, the receiver
direct converter 220 preferably converts the coupledsignal 217 using the same local oscillator signal, VCORX, as the referencedirect converter 238, because thecanceller 230 focuses on the portion of the filtered transmit signal spectrum that is within the receiver bandwidth, as illustrated in FIGS. 2A, 2B and 4. - FIG. 5B illustrates an alternate embodiment of a
transceiver 200′ of FIG. 3 with separate receive and transmitantenna radiators canceller 230. In FIG. 5B, thetransceiver 200′ comprises areceiver bandpass filter 214, a low noise amplifier (LNA) 216, a receiverdirect converter 220, a digital adaptive coherentspectral canceller 230, a matched orchannel filter 250, adirectional coupler 235, atransmitter bandpass filter 218 and atransmitter 280. - In FIG. 5B, the
transmitter 280 receives data and control signals from the digital processor interface. Thetransmitter 280 is coupled to thetransmitter bandpass filter 218, which is coupled to thedirectional coupler 235. Thedirectional coupler 235 is coupled to thetransmitter antenna radiator 122 and thespectral canceller 230. Thereceiver antenna radiator 120 is coupled to thereceiver bandpass filter 214, which is coupled to theLNA 216. TheLNA 216 is coupled to the receiverdirect converter 220, which is coupled to the digital processor interface and thespectral canceller 230. Thespectral canceller 230 is coupled to the processor interface and the matched orchannel filter 250, which is also coupled to the digital processor interface. - Even though there is no
circulator 212 in thetransceiver 200′ of FIG. 5B as in thetransceiver 200 of FIG. 5A, thetransceiver 200′ of FIG. 5B may still experience transmit spectral spillage or leakage into the receive path as the receivingantenna radiator 120 receives the signal energy radiated by the transmittingantenna radiator 122. - In addition to canceling or attenuating leakage from the subject transmitter frequency, the
transceiver 200′ of FIG. 5B can also be configured to cancel interferences from other transmit antennas located in the proximity of the receiver. - FIG. 6 illustrates one embodiment of a digital coherent
spectral canceller 230 within thetransceivers canceller 230 comprises acombiner 232, an adaptive digitaltransversal filter 234, areference bandpass filter 236, a referencedirect converter 238 and a crosscorrelation measurement module 240. - In FIG. 6, the
reference bandpass filter 236 receives a signal from the directional coupler 235 (FIG. 5A) and outputs a signal to the referencedirect converter 238. Thereference converter 238 is coupled to the adaptive digitaltransversal filter 234, the crosscorrelation measurement module 240 and the digital processor interface (FIG. 5A). The adaptive digitaltransversal filter 234 is coupled to thecanceller adaptation module 910 and thecombiner 232, which is coupled to the matched or channel filter 250 (FIG. 5A) and the receiver direct converter 220 (FIG. 5A). As shown in FIG. 5A, thecombiner 232 may also be coupled to the crosscorrelation measurement module 240. The crosscorrelation measurement module 240 is coupled to the matched filter 250 (FIG. 5A) and thecanceller adaptation module 910. - In FIG. 6, the reference
direct converter 238 creates a digitized reference of the spectral energy of thetransmitter 280 within the bandwidth of the receiver to produce an output, Z(n). The output of the referencedirect converter 238, Z(n), comprises an in-phase component, ZI(n), and a quadrature component, ZQ(n), as shown in FIG. 6. The referencedirect converter 238 outputs both the I and Q components of the signal Z(n) to the adaptive digitaltransversal filter 234 and to the crosscorrelation measurement module 240. The crosscorrelation measurement module 240 also receives an I and a Q component of a residue signal R(n) from the matched orchannel filter 250. The adaptive digitaltransversal filter 234 outputs an in-phase component, UI(n), and a quadrature component, UQ(n) (an I and a Q component) of an output signal, U(n), to thecombiner 232. - In FIG. 6, the
combiner 232 preferably uses adigital delay ΔT - In FIG. 6, the in-phase and quadrature component outputs of the
combiner 232 are given as: - Y I(n)=X I(n−Δ)−∪I(n) (1)
- Y Q(n)=X Q(n−Δ)−∪Q(n) (2)
- where the terms are defined as follows:
- XI(n) Digitized output of I-path of
direct converter 220, - XQ(n) Digitized output of Q-path of
direct converter 220, - YI(n) I-path output of
combiner 232, - YQ(n) Q-path output of
combiner 232, - UI(n) I-path output of adaptive digital
transversal filter 234, - UQ(n) Q-path output of adaptive digital
transversal filter 234, - Δ Integer number of sample periods, TS, for delay added to approximately compensate for the difference in delay between the reference and signal paths.
-
- where the terms are defined as follows:
- ZI(n) Digitized output of I-path of the reference
direct converter 238, - ZQ(n) Digitized output of Q-path of the reference
direct converter 238, - WK-I(j) I coefficients of the adaptive digital
transversal filter 234 at the j-th iteration of an adaptation algorithm of thecanceller adaptation module 910, - WK-Q(j) Q coefficients of the adaptive digital
transversal filter 234 at the j-th iteration of an adaptation algorithm of thecanceller adaptation module 910, - K Number of taps (complex coefficients) in the adaptive digital
transversal filter 234. - The
adaptation module 910 adjusts the coefficient vector, W(j), of the adaptive digitaltransversal filter 234 to minimize the transmitter energy within the bandwidth determined by either the bandwidth of thedirect converters bandpass filters - where the terms are defined as follows:
- X(ω) The digitized and filtered output of the receiver
direct converter 220, - Z(ω) The digitized and filtered output of the reference
direct converter 238, - A(ω) The transfer function of the adaptive digital
transversal filter 234, - F(ω) The transfer function of the matched or
channel filter 250, where F(ω)=1 in the alternative embodiment. - ω2 Upper frequency (in radians) over which transmitter spectrum is to be cancelled.
- ω1 Lower frequency (in radians) over which transmitter spectrum is to be cancelled.
- Ts Sampling period of output of receiver
direct converter 220 and reference direct converter 238 (reciprocal of sampling frequency). - Where the coefficients, {[Wk-I(j), Wk-Q(j),], k=1 . . . K}, are selected to minimize E(ω).
- The coefficients associated with the adaptation algorithm used by the adaptive digital
transversal filter 234 which minimize E(ω) are: - W k-I(j+1)=ρW k-I(j)+υC k-I(j) (6)
- W k-Q(j+1)=ρW k-Q(j)+υC k-Q(j) (7)
-
- for k=1 . . . K, with M representing the number of samples in the measurement interval, ρ representing a constant set so that the integrated correlated noise component does not grow with the iterations, and the step size at each iteration.
- Together, FIGS. 9A, 9B,10A and 10B are used to illustrate examples of the cancellation obtainable as a function of the complex coefficients (taps) of the adaptive digital
transversal filter 234, where the bandwidth of F (ω) is one quarter of the sampling frequency. In FIGS. 9A-10B, the horizontal axis represents frequency× 1000/sample rate. The vertical axis represents amplitude in decibels (dB). - FIG. 9A illustrates exemplary effective baseband responses (transfer functions)430, 432 or one embodiment of the receive
filter 214 and the transmitreference filter 236 of FIG. 5A, where the receiver andreference filters baseband responses baseband response filter 214 and/or the transmitreference filter 236. - FIG. 9B illustrates 10 log10 |Y(ω)2/|X(ω)2, cancellation ratios in dB of the spectral leakage from the
transmitter 280 due to the digital coherent spectral 230 of FIG. 6. Curves 434-440 represent a different number of taps for the digital coherent spectral 230 of FIG. 6, where the receiver andreference filters spectral canceller 230 reflects cancellation over the bandwidth of ω1=0 to ω2=250. In FIG. 9B theline 434 represents the amount of cancellation achieved with an adaptive digitaltransversal filter 234 of one tap, which does not compensate (equalize) for the frequency dependent mismatch characteristics between the two paths. As shown in FIG. 9B, if the adaptive digitaltransversal filter 234 uses an increased number of taps, the cancellation performance is significantly improved. - The filtering of the reference transmitter output signal with a
reference filter 236 having substantially the same design as the receiver filter 214 (FIG. 9A) (i.e., the transfer functions HRX (ω) and HREF (ω) in FIG. 5A are substantially similar) presents at least two advantages. First, it may reduce dynamic range requirements by providing high attenuation to signals outside the receiver band so that the reference direct converter 238 (FIG. 5A) only digitizes the signal within the spectral sidelobes of the transmit signal that fall within the receiver bandwidth. This attenuation may significantly reduce the resolution and linearity requirements of the referencedirect converter 238 because the sidelobes within the receiver filter's bandwidth will normally be suppressed more than 40 dB below the peak of the transmitter signal. - Second, using a
reference bandpass filter 236 with the same design (i.e., transfer function) as thereceiver filter 214 associated with theduplexer 210 may also increase the amount of suppression because thefilters direct converters transversal equalizer 234 may advantageously compensate for any small differences in the transfer functions using a smaller number of taps than required if HRX (ω) and HREF (ω) in FIG. 5A were not substantially similar. - FIG. 10A illustrates exemplary
cancellation ratio responses receiver filter 214 and the transmitreference filter 236 of FIG. 5A, where the receiver and reference filters, HRX (ω) and HREF (ω), are Chebyshev bandpass filters designed for different bandwidths and with different pass band ripples, such as 0.5 dB and 1.5 dB respectively. Thetransfer functions transfer functions baseband responses baseband response filter 214 and/or the transmitreference filter 236. - FIG. 10B illustrates 10 log10|Y(ω)|2/X(ω)|2, cancellation ratios in dB of the spectral leakage from the
transmitter 280 due to the digital coherent spectral 230 of FIG. 6. Curves 446-452 represent a different number of taps for the digitalspectral canceller 230 of FIG. 6, where the receiver andreference filters transfer functions transfer functions transversal filter 234 to achieve substantially the same level of cancellation performance as the performance shown in FIG. 9B. - FIG. 7 illustrates one embodiment of a
direct converter 220 within thetransceiver direct converter 238 is substantially similar to the structure of the receiverdirect converter 220. In FIG. 7, thedirect converter 220 comprises anLNA 202, adivider 205, a translating delta-sigma modulator 213 for the in-phase path, a translating delta-sigma modulator 211 for the quadrature path, a receiver frequency synthesizer andclock generator 500, and I & Q decimation filters 300. In one embodiment, theLNA 202 and theLNA 216 of FIG. 5A are two separate components. In another embodiment, theLNA 202 and theLNA 216 represent a single LNA component. Thedivider 205 may be implemented with a variety of active elements and/or passive elements. Thedivider 205 may be implemented as a splitter. Alternatively, the output of theLNA 202 can be coupled directly to both of the translating delta-sigma modulators - The
LNA 202 of thedirect converter 220 and/or theLNA 216 receives anRF signal 201 from the receiver bandpass filter 214 (FIG. 5A). TheLNA 202 is coupled to thedivider 205, which is coupled to both translating delta-sigma modulators sigma modulators clock generator 500. The receiver synthesizer andclock generator 500 is coupled to the reference direct converter 238 (FIG. 5A) and the digital processor interface (FIG. 5A). The decimation filters 300 are coupled to the combiner 232 (FIG. 5A). - In FIG. 7, the
LNA 202 amplifies theincoming signal 201 while avoiding the addition of excessive noise and distortion. Thedivider 205 separates the amplified signal into in-phase and the quadrature components and outputs the in-phase and quadrature components to the in-phase andquadrature modulators modulators modulators decimation filter 300. In one embodiment, the digital data output of each of themodulators - The
decimation filter 300 performs digital filtering and decimation to produce digital words at the rate of a clock, CLK. In FIG. 7, thedecimation filter 300 converts the stream of 1-bit digital words into a stream of N-bit digital words that comprise conventional binary representations of the signals. In one embodiment, thedecimation filter 300 produces a clock signal, CLK_M, which indicates the rate at which the binary representations are created and is used to transfer the filtered output. The rate at which the words produced by thedecimation filter 300 is determined by the oversampling ratio. For an oversampling ratio of M, the output clock rate is 1/M times the rate of the incoming data, or alternatively stated, for the embodiment of FIG. 13, the output clock rate is equal to fRX/M where fRX is the rate of the translation and conversion clock, CLK. - The
decimator filter 300 also attenuates unwanted signals and noise sources outside of the bandwidth of interest while preserving the signals of interest. The desired filter characteristics of thedecimation filter 300 are selected on the basis of the characteristics of the receive signal as well as the conversion clock rate, fRX. In one embodiment, thedecimation filter 300 has a programmable characteristic which can be modified to accommodate a variety of different waveforms. Thedecimation filer 300 is preferably implemented with a finite impulse response (FIR) filter whose characteristics are modified by changing the value of the filter coefficients. The value of the filter coefficients of a digital filter can be readily changed via software making such modifications practical. - FIG. 8 illustrates another embodiment of a
direct converter 800 within thetransceiver direct converter 800 comprises anLNA 802, a pair ofamps balanced mixers amps digital converters quadrature hybrid module 805 and a receiver frequency synthesizer and VCO 816. In one embodiment, theLNA 802 and theLNA 216 of FIG. 5A are two separate components In another embodiment, theLNA 802 and theLNA 216 represent a single LNA component. - The
LNA 802 and/or theLNA 216 receives the RF signal 201 from the receiver bandpass filter 214 (FIG. 5A). TheLNA 202 is coupled to thedivider 804, which is coupled to bothamps amp 808 is coupled to thebalanced mixer 810, which is coupled to thequadrature hybrid module 805 and the low pass filter/amp 812. The low pass filter/amp 812 is coupled to the analog-digital converter 814, which is coupled to the analogdigital convert 815, the combiner 232 (FIG. 5A) and the digital processor interface (FIG. 5A). Theamp 809 is coupled to thebalanced mixer 811, which is coupled to thequadrature hybrid module 805 and the low pass filter/amp 813. The low pass filter/amp 813 is coupled to the analog-digital converter 815, which is coupled to the analogdigital converter 814, the combiner 232 (FIG. 5A) and the digital processor interface (FIG. 5A). - FIG. 11 illustrates another embodiment of a
direct converter 220′ within thetransceiver direct converter 220′ comprises adivider 205, a translating delta-sigma modulator 213 for the in-phase path, a translating delta-sigma modulator 211 for the quadrature path, aclock generator 504, an I & Q decimation filters 300 and an I & Q gain quadrature and offsetcorrection module 410. In one embodiment, thedirect converter 220′ of FIG. 11 may include an input LNA coupled to the RF input of thedivider 205. - In FIG. 11, the
divider 205 receives an RF input from the LNA 216 (FIG. 5A). Thedivider 205 is coupled to both translating delta-sigma modulators sigma modulators clock generator 504. Theclock generator 504 is coupled to the reference direct converter 238 (FIG. 5A) and the digital processor interface (FIG. 5A). The decimation filters 300 are coupled to the I & Q gain quadrature and offsetcorrection module 410, which is coupled to the combiner 232 (FIG. 5A). - FIG. 12 illustrates another embodiment of a
direct converter 800′ within thetransceiver direct converter 800′ comprises adivider 804, a pair ofamps balanced mixers amps digital converters quadrature hybrid module 805 and an I & Q gain, quadrature and offsetcorrection module 850. In one embodiment, thedirect converter 800′ of FIG. 12 may include an input LNA coupled to the RF input of thedivider 205. - In FIG. 12, the
divider 804 receives an RF input from the LNA 216 (FIG. 5A). Thedivider 804 is coupled to bothamps amp 808 is coupled to thebalanced mixer 810, which is coupled to thequadrature hybrid module 805 and the low pass filter/amp 812. The loss pass filter/amp 812 is coupled to the analog-digital converter 814, which is coupled to the analogdigital converter 815, the I & Q gain, quadrature and offsetcorrection module 850 and the digital processor interface (FIG. 5A). Theamp 809 is coupled to thebalanced mixer 811, which is coupled to thequadrature hybrid module 805 and the low pass filter/amp 813. The low pass filter/amp 813 is coupled to the analog-digital converter 815, which is coupled to the analogdigital converter 814, the I & Q gain, quadrature and offsetcorrection module 850 and the digital processor interface (FIG. 5A). - FIG. 13 illustrates one embodiment of a translating delta-
sigma modulator 211 within thedirect converters modulator 213 is substantially similar to themodulator 211, only themodulator 211 will be described herein. In FIG. 13, themodulator complementary amplifier 310, aswitch 312, aloop amplifier 314, a one-bit digital-to-analog converter 316, aloop filter 318 and an edge-triggeredcomparator 320. - In FIG. 13, the
output 206 of the divider 205 (FIGS. 7 or 11) is input into thecomplementary amplifier 310, which is coupled to theswitch 312. Theswitch 312 is coupled to theloop amplifier 314. Theloop amplifier 314 is coupled to the digital-to-analog converter 316 and theloop filter 318, which is coupled to thecomparator 320. The CLK signal is coupled to theswitch 312 and the edge-triggeredcomparator 320. - The
modulator 211 translates the received I & Q signal components by an amount equal to the frequency of the CLK signal, fRX, to the baseband of the received signal and digitizes the I & Q signal components. In FIG. 13, thecomplementary amplifier 310 receives the modulated RF carrier signal. At a non-inverting output, thecomplementary amplifier 310 produces a voltage that is G times the voltage at the input to thecomplementary amplifier 310. At an inverting output, thecomplementary amplifier 310 produces a voltage that is −G times the voltage at the input to thecomplementary amplifier 310. The inverting and non-inverting outputs of thecomplementary amplifier 310 are coupled to two input ports of aswitch 312. The control port of theswitch 312 determines which input port of theswitch 312 is coupled to the output port of theamplifier 310 and is driven by the conversion clock, CLK, such that the output port of theswitch 312 is alternately coupled to the inverting and non-inverting outputs of thecomplementary amplifier 310. - Together, the
complementary amplifier 310 and theswitch 312 perform the functions of a commutator which inverts the polarity of the modulated RF carrier signal on every half cycle of the conversion clock, CLK. If the frequency of the conversion clock CLK is chosen to be approximately equal to the carrier frequency of the modulated RF carrier signal, effectively, the commutator translates the modulation of the carrier signal down to D.C. centered or frequency offset baseband. In addition to the low frequency signal components, high frequency signal components are also generated by the commutator. However, the high frequency components are attenuated by the delta-sigma modulator 211 and further filtering. In one embodiment, the frequency of the conversion clock CLK is programmable to permit the translation of a variety of waveforms over a range of center frequencies. - In one embodiment, the commutator comprised of the
complementary amplifier 310 and theswitch 312 is not a conventional downconverter. The mathematical paradigm for a conventional downconverter is multiplication by a sinusoidal signal. Practical implementations of conventional downconverters (such as circuits employing using diode rings or Gilbert multiplier circuits) are incapable of realizing this mathematical paradigm without introduction of distortion and feed-through effects that result in the creation of undesired spurious signals. - In contrast, the mathematical paradigm of the commutator is that of alternately multiplying the input signal by +1 or −1 on opposite half cycles of a clock signal. Practical implementations of the commutator employing a fast switch behave more closely to this mathematical paradigm, thus avoiding the production of non-linear components of the signal in the baseband signal in comparison to a conventional down converter.
- The output of the
switch 312 is coupled to the input of the core delta-sigma modulator, which comprises aloop amplifier 314, aloop filter 318, an edge-triggeredcomparator 320 and a one-bit digital-to-analog (D/A)converter 316. In the preferred embodiment, the core delta-sigma modulator is operated at the same frequency as the commutator. Use of a conversion clock operating at or near the carrier frequency provides a significant oversampling ratio in typical embodiments and, hence, leads to high resolution, high dynamic range performance according to well-known principles of delta-sigma conversion. - In FIG. 13, the output of the
loop amplifier 314 is the difference between the voltage coupled to its non-inverting input port and its inverting input port times a voltage gain, AV, where the voltage gain is typically a large positive constant. Theloop filter 318 is typically an analog low pass filter but can be embodied in other forms. In one embodiment, theloop amplifier 314 andloop filter 318 act as an integrator. When the voltage value at the signal input to the edge-triggeredcomparator 320 is greater than a predetermined threshold value at the time the conversion clock transitions, the output is alogic value 1. When the voltage value at the signal input to the edge-triggeredcomparator 320 is less than the predetermined threshold value at the time the conversion clock transitions, the output is alogic value 0. The output of the edge-triggeredcomparator 320 is coupled to the input of the one-bit digital-to-analog converter 316. The one-bit digital-to-analog converter 316 produces one of two analog levels at its output depending upon the digital logic value applied to its input. The output of the one-bit digital-to-analog converter 316 is coupled to the inverting input of the loop amplified 314. - The core delta-
sigma modulator - Because metal-oxide semiconductor (MOS) technology lends itself inherently to implementing discrete-time filters based on capacitor ratios, prior art systems use switched-capacitor technology to implement delta-sigma modulators. Inherently, switched-capacitor filters cause aliasing and hence, additional interference to the system. In addition, because MOS switched-capacitor circuits must be operated at a much lower oversampling ratio, they do not have as much resolution for any given order of the delta-sigma modulator, compared to one embodiment of the
modulator 211 of the present invention. In order to gain resolution, prior art systems typically use higher order loop filters which are only conditionally stable. As the order of the delta-sigma modulator is increased, the implementation of a stable loop that is capable of operating at high clock frequencies becomes more difficult. - In contrast, one embodiment of the
modulator 211 comprises a continuous-time filter for theloop filter 318. As noted above, one embodiment of the delta-sigma modulator 211 operates at or near the carrier frequency. Due to the use of a high frequency clock, the use of higher order filtering is not needed to achieve a high degree of resolution. Therefore, the use of a lower order, continuous-time filter is practical in conjunction with one embodiment of themodulators - Many modem delta-sigma converters are currently available that are implemented in silicon metal oxide semiconductor (MOS) technology. Typically such designs use switched capacitor techniques to sample the incoming signal for conversion. However, circuits capable of processing high frequency input signals, such as those formed from silicon bipolar, silicon germanium (SiGe), or gallium arsenide (GaAs) technologies, can use current steering architectures in order to increase system efficiencies.
- FIGS.14A-14E are spectral plots used to illustrate the operation of various embodiments of the
transceiver 200 of FIG. 1. An understanding of the desired characteristics of thedecimation filter 300 can be understood with reference to FIG. 14A where the vertical axis represents energy such as in units of decibels and the horizontal axis represents frequency such as in units of Gigahertz. FIG. 14A is a spectral plot showing receivedsignal energies signal energies - FIG. 14B represents the corresponding output of the switch313 (excluding noise) of FIG. 13, when the spectrum shown in FIG. 14 is applied thereto. For example, in a typical embodiment, the frequency fRX is equal to 1851.4 MHz and the frequencies fc1, fc2 and fc3 are 1851, 1851.6 and 1852.2 MHz, respectively. Each of the
signal energies signal energies energies signal energy 336 has been translated to the negative portion of the frequency axis. - A dashed
line 342 of FIG. 14C represents the transfer curve of thedecimation filter 300 in one embodiment. In the embodiment of FIG. 14C, the lowpass decimation filter 300 passes all threesignal energies signal energies noise density curve 343 around zero frequency) can be reduced by follow-on filtering, for example, matched filtering. The spectralnoise density curve 343 of FIG. 14C shows that the spectral noise density level increases (primarily due to quantization noise of the delta-sigma converter) as the frequency increases, expect for 1/f increase near zero frequency. As a result, the noise level within the bandwidth ofsignal energy 340 is greater than that forsignal energies decimation filter 300 is implemented with low pass filtering and equivalent bandpass filtering is implemented in the following matched filter. - In an alternative embodiment, the
decimation filter 300 is more frequency selective such that only one of the signal energies (such as might be produced by a single transmitting unit) is passed without substantial attenuation. For example, in FIG. 14D, the dashedline 344 shows such adecimation filter 300 transfer characteristic. As shown in FIG. 14D, in the alternative embodiment, only thesignal energy 338 is efficiently passed through thedecimation filter 300. - In yet another embodiment, the down-converted waveform is centered about the D.C., i.e., the waveform has zero frequency offset, as shown in FIG. 14E. Conversion to D.C. centered baseband has the benefit of achieving higher resolution for a given clock rate which can be a particular benefit for wide band signals where the effects of quantization noise should be minimized. The effects of 1/f noise are less pronounced in a wide band system and can be filtered with a notch filter at zero frequency without significantly degrading the performance. A dashed
line 346 of FIG. 14E represents the transfer curve of thedecimation filter 300 in one such embodiment. More information concerning the design of decimation filters can be found in Multi-Rate Digital Signal Processing, Prentice-Hall, Inc., Englewood Cliffs, N.J., 1983 by R. E. Crochiere and L. R. Rabiner. - It is advantageous for the receiver106 (FIG. 1) to operate in accordance with more than one communication protocol. For example, the receiver can operate in a narrow band time division multiple access (TDMA) system such as Global System for Mobile Communication (GSM) or a wide band code division multiple access (CDMA) system such as defined in the Telephone Industry Association, Electronic Industry Association (TIA/EIA) interim standard entitled “Mobile Station—Base Station Capability Standard for Dual-Mode Wide band Spread Spectrum Cellular System,” TIA/EIA/IS-95.
- During TDMA operation, the
decimation filter 300 can take on a narrow band transfer characteristic, shown by dashedline 344. Alternatively, during CDMA operation, the offset and bandwidth of this filter, shown byline 342, may be increased according to well known principles of digital filtering and signal reception. Alternatively, a single wide band, low-pass decimation filter could be utilized and the programmable bandwidth implemented in the following matched filtering. - I&Q Gain Quadrature and Offset Correction
- In FIG. 11, the output of the
decimation filter 300 is the input into the I & Q gain quadrature and offsetcorrection component 410. FIG. 15 is a block diagram showing one embodiment of thecomponent 410. In FIG. 15, the clock and data output of the decimation filters 300 are coupled to acalibration circuit 350. Thecalibration circuit 350 adjusts the relative gain and phase so that the in-phase and quadrature signal paths are balanced with respect to each other. In order to avoid introduction of distortion into the signals, it is important that the relative gain and phase of the in-phase and quadrature signal paths are the same. - One advantage of a digital signal processing architecture shown in FIG. 15 is that the parameters can be controlled in the digital circuit elements more easily than in analog circuit elements. Typically, unbalances originate from the differences in gain between the I and Q channels and errors in the relative 90° phase shift between the I and Q channels. Additionally, any differences in the DC offsets can be calibrated out. Additional information concerning accomplishment of calibration can be found in U.S. Pat. No. 5,422,889 entitled “OFFSET CORRECTION CIRCUIT,” and in U.S. Pat. No. 5,604,929 entitled “SYSTEM FOR CORRECTING QUADRATURE GAIN IN-PHASE ERROR IN A DIRECT CONVERSION SINGLE-SIDE BAND RECEIVER INDEPENDENT OF THE CHARACTERISTICS OF THE MODULATED SIGNAL.”
- In FIG. 15, the output of the
calibration circuit 350 is coupled to the input of asampling rate converter 352. Thesampling rate converter 352 converts and synchronizes the data rate of the signal to the rate of an external clock, CLKwaveform. In one embodiment, this function is accomplished with a linear or higher order interpolation method such as the one described in “Advanced Digital Signal Processing” by J. G. Proakis, et al., and McMillan Publishing Co. - In FIG. 15, the output of the
sampling rate converter 352 is coupled to the input of afrequency translator 354. In one embodiment, thefrequency translator 354 is used to translate the center frequency of the signal of interest to a D.C. centered baseband. Thefrequency translator 354 multiplies the signal at the output of thesampling rate converter 352 with a digital representation of a sinusoidal signal having a frequency equal to the center of the frequency of signal of interest. The advantage of frequency translation is that it allows the matchedfilter 356 for the signal to be implemented as a low pass filter and provides the baseband I and Q inputs required for the digital demodulator input. For the situation shown in FIG. 14E where there is only one signal of interest and it has zero offset, thefrequency translator 354 is preferably not used. - When the
frequency translator 354 is utilized, and the output of the matched orchannel filter 250 is utilized as an input to the crosscorrelation measurement module 240 as shown in FIG. 5A, then the input Z(n) to the crosscorrelation measurement module 240 must also undergo the same frequency translation. The translation can be accomplished by including a frequency translator in the crosscorrelation measurement module 240. - In FIG. 15, the output of the
frequency translator 354 is coupled to alow pass filter 356 which can operate as a signal matched filter. Thelow pass filter 356 is also used to reject interference outside the bandwidth of interest. The output of thelow pass filter 356 provides a digital I and Q signal input to the digital demodulator that is synchronized with the digital demodulator clock—CLKWAVEFORM. - FIG. 16 is a block diagram showing one embodiment of the
clock generator 500 of FIG. 7. In FIG. 16, afrequency synthesizer 360 produces an analog waveform at twice the rate of the conversion clock, CLK. The output of thefrequency synthesizer 360 is coupled to the input of a limitingamplifier 362. In this embodiment, the positive going zero crossing of the signal output by thefrequency synthesizer 360 is compared to a threshold by the limitingamplifier 362. When the threshold is chosen appropriately, the limitingamplifier 362 produces a waveform with digital logic values at the same frequency as that of the output from thefrequency synthesizer 360 and having a 50% duty cycle (i.e., the duration of the logic “1” pulse is the same as the duration of the logic “0” pulse). - In FIG. 16, the limiting
amplifier 362 drives a master slave flip-flop 376 comprising amaster latch 364 and aslave latch 368. The master-slave flip-flop 376 is configured in a divide-by-two configuration. In this configuration, aQ output 366 and a {overscore (Q)}output 372 of the flip-flop 364 are connected to the D and {overscore (D)} inputs of the flip-flop 368, respectively, and aQ output 370 and a {overscore (Q)}output 374 of the flip-flop 368 are connected to the {overscore (D)} and D inputs of the flip-flop 364, respectively. When the master-slave flip-flop is connected in this manner, the fourlatch outputs output 366 and output 370) can be used as I_CLK and Q_CLK, respectively. Although the implementation of FIG. 16 is included explicitly herein for illustration purposes, a variety of other means (such as a ring oscillator) can be used to generate a clock signal in accordance with the present invention. - FIG. 17 illustrates another embodiment of a translating delta-sigma modulator380 which employs double-sampling (i.e., samples on both edges of a clock signal). The delta-sigma modulator 380 of FIG. 17 operates under some of the same principles as the single sampled architecture shown in FIG. 13 while doubling the sample rate, thereby relaxing the speed requirements for circuitry by a factor of two. The delta-sigma modulator 380 can be used as within the architecture shown in FIG. 7 as the transmitting delta-
sigma modulators - In FIG. 17,
complementary amplifier 382 receives the digitally modulated RF signal centered about the carrier frequency. At a non-inverting output, thecomplementary amplifier 382 produces a voltage that is G times the voltage at the input to thecomplementary amplifier 382. At an inverting output, thecomplementary amplifier 382 produces a voltage that is −G times the voltage at the input to thecomplementary amplifier 382. The inverting and non-inverting outputs of thecomplementary amplifier 382 are coupled to two input ports of aswitch 384. The control port of theswitch 384 determines which input port is coupled to the output port and is driven by the conversion clock, CLK, such that the output port of theswitch 384 is alternately coupled to the inverting and non-inverting outputs of thecomplementary amplifier 382. - Together, the
complementary amplifier 382 and theswitch 384 perform the functions of a commutator as explained more fully above with reference to FIG. 13. The output of theswitch 384 is coupled to the input of the core double-sampling delta-sigma modulator. The core double-sampling delta-sigma modulator is comprised of acombiner 388, aloop amplifier 390, aloop filter 392, an even-phase edge-triggeredcomparator 394A, an odd-phase edge-triggeredcomparator 394B, an even-phase digital-to-analog converter 396A and an odd-phase digital-to-analog converter 396B. - The output of the
switch 384 is coupled to the non-inverting input of theloop amplifier 390. The output of theloop amplifier 390 is the difference between the voltage coupled to its non-inverting input port and its inverting input port times a voltage gain Av where the voltage gain is typically a large positive constant. The output of theloop amplifier 390 is coupled to the input of theloop filter 392. In a preferred embodiment, theloop filter 392 is an analog low pass filter but can be embodied in other forms. In one embodiment, theloop amplifier 390 andloop filter 392 act as an integrator. - The output of the
loop filter 392 is coupled to the input of the even-phase edge-triggeredcomparator 394A and also to the input of the odd-phase edge-triggeredcomparator 394B. The clock inputs of the even-phase edge-triggeredcomparator 394A and the odd-phase edge-triggeredcomparator 394B are coupled to the conversion clock, CLK. The even-phase edge-triggeredcomparator 394A and odd-phase edge-triggeredcomparator 394B are clocked using opposite edges of the comparison clock, CLK. For example, in one embodiment, the even-phase edge-triggeredcomparator 394A performs a comparison on the rising edge of the comparison clock, CLK, and the odd-phase edge-triggeredcomparator 394B performs a comparison on the falling edge of the comparison clock, CLK. - The logic values output by the even-phase edge-triggered
comparator 394A and the odd-phase edge triggeredcomparator 394B are coupled to the input of the digital-to-analog converter 396A and the digital-to-analog converter 396B, respectively. The outputs of the digital-to-analog converter 396A and digital-to-analog converter 396B are combined through thecombiner 388 and drive the inverting input of theloop amplifier 390. In one embodiment, thecombiner 388 simply adds the two values together. In another embodiment, thecombiner 388 time-division multiplexes the values into the loop. One useful attribute of the first embodiment of thecombiner 388 is that linearity can be achieved without tight matching between the tight-to-analog converter 396A and the digital-to-analog converter 396B since their respective outputs are effectively averaged before being presented to the loop amplifier. - In one embodiment, the outputs of edge-triggered
comparator 394A and edge-triggeredcomparator 394B are also coupled to thedecimation filter 300 in a similar manner to the single-sampled case. In such an embodiment, typically the architecture of thedecimation filter 300 is appropriately modified to accommodate processing the samples in the form of two bit serial words instead of a single high speed serial bit stream. - Due to the continuous time nature of the direct converter embodiments shown in FIGS. 7, 13 and17, the direct converter embodiments are not limited in dynamic range in contrast to conventional multistage down converters. Thus, there is no need to incorporate automatic gain control into the front end of the receiver. For example, in FIG. 5A, the amplitude of the incoming waveform applied to the receiver
direct converter 220 is in fixed proportion of an amplitude of a signal received by the receiveantenna radiator 102 because no automatic gain control mechanism is included. As noted above, automatic gain control used to extend the dynamic range of a receiver may be undesirable for spectrally crowded applications such as cellular communications because the automatic gain control may make the receiver sensitivity dependent upon signals and interference that are outside the signal channel. For example, it is possible for a strong signal in an adjacent channel to capture the receiver front end and desensitize the receiver such that a weak signal in the channel of interest is undetectable. This is particularly harmful in a base station receiver where the receiver receives incoming signals from multiple remote units. Furthermore, the use of automatic gain control will likely require the digital coherent canceller to track the gain changes which will introduce errors and noise. - The digital decimation filtering of the receiver direct converter220 (FIGS. 5A, 7 and 8), the reference direct converter 238 (FIG. 5A), and the matched or channel filter 250 (FIG. 5A) with low side lobes may significantly suppress the transmit signal outside of the desired signal bandwidth. In a preferred implementation, these digital decimation filters 220, 238, 250 have approximately 90 dB attenuation. This attenuation advantageously reduces the amount of filtering provided by the
duplexer receiver filter 214. - In one embodiment, the reference
direct converter 238 and/or the receiverdirect converter 220 have a sampling rate approximately equal to that of the carrier frequency of interest. This may significantly reduce the requirements on theduplexer receiver filter 214 and thereference bandpass filter 236 to provide attenuation at the aliasing frequency (i.e., at half the RF frequency). - The invention may be embodied in other specific forms without departing from its spirit or essential characteristics. The described embodiment is to be considered in all respects only as illustrative and not restrictive and the scope of the invention is, therefore, indcated by the appended claims rather than the foregoing description. All changes which come within the meaning and range of equivalency of the claims are to be embraced within their scope.
Claims (12)
1. A transceiver comprising:
a receiver direct converter translating a received signal to a baseband of the received signal and digitizing the translated, received signal;
an adaptive canceller comprising a reference direct converter, the reference direct converter outputting a digitized transmit signal reference of a spectral energy of a transmitter within the bandwidth of a receiver; and
a matched filter, wherein the receiver direct converter, the reference direct converter, and the matched filter suppress the spectral energy of the transmitter within the bandwidth of the receiver.
2. The transceiver of claim 1 , wherein the transceiver is a full duplex transceiver.
3. The transceiver of claim 1 , further comprising a transmit and receive antenna radiator.
4. The transceiver of claim 1 , further comprising a transmit antenna radiator and a receive antenna radiator.
5. The transceiver of claim 1 , where the receiver direct converter, the reference direct converter, and the matched filter have approximately 90 dB attenuation.
6. The transceiver of claim 1 , wherein the receiver direct converter has a sampling rate approximately equal to that of the carrier frequency of interest.
7. The transceiver of claim 1 , wherein the reference direct converter has a sampling rate approximately equal to that of the carrier frequency of interest.
8. The transceiver of claim 1 , wherein the canceller further comprises an adaptive digital transversal filter adapted to align an amplitude and a phase of the digitized transmit signal reference in a reference path with a transmit signal in a leakage receiver path, the adaptive digital transversal filter outputting an compensated digitized transmit signal reference.
9. The transceiver of claim 1 , wherein the transceiver is adapted to cancel interference from other co-sited transmit antennas.
10. A method of attenuating a transmitter signal spectrum within a bandwidth of a receiver, the method comprising:
digitizing a received signal which is corrupted by components of a transmit signal;
creating a digitized reference transmit signal of the transmit signal within the bandwidth of the receiver;
aligning the digitized reference transmit signal in amplitude, phase and time delay with the digitized received signal;
subtracting the digitized reference transmit signal from the digitized received signal to form a residue; and
suppressing a transmitter spectral signal power of the residue within the bandwidth of the receiver.
11. The method of claim 10 , further comprising adjusting the transmit signal based on the residue determined by subtracting the digitized reference transmit signal from the digitized received signal.
12. A transceiver comprising:
a duplexer coupled to an antenna;
a receiver receiving a first signal from the duplexer;
a transmitter sending a second signal to the duplexer; and
an adaptive, digital, coherent spectral canceller coupled to the receiver and the transmitter, the canceller attenuating a signal spectrum leakage of the second signal within a bandwidth of the first signal.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US10/758,355 US20040151238A1 (en) | 2000-01-18 | 2004-01-15 | Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth |
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US09/487,396 US6704349B1 (en) | 2000-01-18 | 2000-01-18 | Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth |
US10/758,355 US20040151238A1 (en) | 2000-01-18 | 2004-01-15 | Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth |
Related Parent Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US09/487,396 Continuation US6704349B1 (en) | 2000-01-18 | 2000-01-18 | Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth |
Publications (1)
Publication Number | Publication Date |
---|---|
US20040151238A1 true US20040151238A1 (en) | 2004-08-05 |
Family
ID=23935562
Family Applications (2)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US09/487,396 Expired - Lifetime US6704349B1 (en) | 2000-01-18 | 2000-01-18 | Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth |
US10/758,355 Abandoned US20040151238A1 (en) | 2000-01-18 | 2004-01-15 | Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth |
Family Applications Before (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US09/487,396 Expired - Lifetime US6704349B1 (en) | 2000-01-18 | 2000-01-18 | Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth |
Country Status (5)
Country | Link |
---|---|
US (2) | US6704349B1 (en) |
EP (1) | EP1249077A2 (en) |
JP (1) | JP2003520549A (en) |
AU (1) | AU2001231011A1 (en) |
WO (1) | WO2001054290A2 (en) |
Cited By (53)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20050069394A1 (en) * | 2003-09-26 | 2005-03-31 | Dyer Kenneth C. | Method, device and system for output impedance calibration that invariably maximizes hybrid performance |
US20050135461A1 (en) * | 2003-12-22 | 2005-06-23 | Nokia Corporation | Deconvolution searcher for wireless communication system |
US20060025072A1 (en) * | 2004-07-29 | 2006-02-02 | Lucent Technologies, Inc. | Extending wireless communication RF coverage inside building |
US20070015468A1 (en) * | 2005-04-14 | 2007-01-18 | Kouki Ammar B | Balanced active and passive duplexers |
US20070104254A1 (en) * | 2003-11-24 | 2007-05-10 | Bottomley Gregory E | Multi-Transmitter Interference Suppression Using Code-Specific Combining |
US20070280385A1 (en) * | 2006-05-30 | 2007-12-06 | Benq Corporation | Method and apparatus of receiving signals and wireless multimode wideband receiver |
US20080062029A1 (en) * | 2006-09-12 | 2008-03-13 | Xu Changqing | Apparatus and method for demodulating a modulated signal |
US20080130719A1 (en) * | 2006-12-05 | 2008-06-05 | Bottomley Gregory E | Method and Apparatus for Suppressing Interference Based on Channelization Code Power Estimation with Bias Removal |
US20090016545A1 (en) * | 2006-08-23 | 2009-01-15 | Quellan, Inc. | Pre-configuration and control of radio frequency noise cancellation |
US20090116455A1 (en) * | 2007-11-06 | 2009-05-07 | Bottomley Gregory E | Method and Apparatus for Code Power Parameter Estimation for Received Signal Processing |
US20090115579A1 (en) * | 2007-11-06 | 2009-05-07 | Microelectronics Technology Inc. | Signal processing apparatus for receiving rfid signal and method thereof |
US20100048146A1 (en) * | 2007-01-30 | 2010-02-25 | Crestcom, Inc. | Transceiver with Compensation for Transmit Signal Leakage and Method Therefor |
US20100150033A1 (en) * | 2008-12-16 | 2010-06-17 | General Electric Company | Software radio frequency canceller |
US20100150032A1 (en) * | 2008-12-12 | 2010-06-17 | General Electric Company | Software radio frequency canceller |
US20110228828A1 (en) * | 2008-11-21 | 2011-09-22 | Wei Wang | Method and device for cancelling transmitter interference in transceiver, and transceiver |
US20110243202A1 (en) * | 2010-04-01 | 2011-10-06 | Ismail Lakkis | Cancellation System for Millimeter-Wave Radar |
CN102771054A (en) * | 2010-02-11 | 2012-11-07 | 联发科技(新加坡)私人有限公司 | Integrated circuits, communication units and methods of cancellation of intermodulation distortion |
US8422540B1 (en) * | 2012-06-21 | 2013-04-16 | CBF Networks, Inc. | Intelligent backhaul radio with zero division duplexing |
US8467363B2 (en) | 2011-08-17 | 2013-06-18 | CBF Networks, Inc. | Intelligent backhaul radio and antenna system |
US8502733B1 (en) | 2012-02-10 | 2013-08-06 | CBF Networks, Inc. | Transmit co-channel spectrum sharing |
USD704174S1 (en) | 2012-08-14 | 2014-05-06 | CBF Networks, Inc. | Intelligent backhaul radio with symmetric wing radome |
US20140161005A1 (en) * | 2012-11-13 | 2014-06-12 | Stephane Laurent-Michel | Method and system for signal dynamic range improvement for frequency-division duplex communication systems |
US8761100B2 (en) | 2011-10-11 | 2014-06-24 | CBF Networks, Inc. | Intelligent backhaul system |
US8811365B2 (en) | 2011-08-17 | 2014-08-19 | CBF Networks, Inc. | Intelligent backhaul radio |
US20140247757A1 (en) * | 2013-03-01 | 2014-09-04 | Qualcomm Incorporated | Multi-tap adaptive filter for transmit signal leakage cancellation |
US8872715B2 (en) | 2011-08-17 | 2014-10-28 | CBF Networks, Inc. | Backhaul radio with a substrate tab-fed antenna assembly |
US8942216B2 (en) | 2012-04-16 | 2015-01-27 | CBF Networks, Inc. | Hybrid band intelligent backhaul radio |
US8982772B2 (en) | 2011-08-17 | 2015-03-17 | CBF Networks, Inc. | Radio transceiver with improved radar detection |
US8989762B1 (en) | 2013-12-05 | 2015-03-24 | CBF Networks, Inc. | Advanced backhaul services |
US9049611B2 (en) | 2011-08-17 | 2015-06-02 | CBF Networks, Inc. | Backhaul radio with extreme interference protection |
CN104967461A (en) * | 2014-03-31 | 2015-10-07 | 英特尔Ip公司 | Self-interference rejection based on correlation |
US9178559B2 (en) | 2008-12-12 | 2015-11-03 | St Ericsson Sa | Method and system of calibration of a second order intermodulation intercept point of a radio transceiver |
US9474080B2 (en) | 2011-08-17 | 2016-10-18 | CBF Networks, Inc. | Full duplex backhaul radio with interference measurement during a blanking interval |
US9603155B2 (en) * | 2015-07-31 | 2017-03-21 | Corning Optical Communications Wireless Ltd | Reducing leaked downlink interference signals in a remote unit uplink path(s) in a distributed antenna system (DAS) |
US9713019B2 (en) | 2011-08-17 | 2017-07-18 | CBF Networks, Inc. | Self organizing backhaul radio |
US20170264327A1 (en) * | 2016-02-23 | 2017-09-14 | Resonant Sciences, LLC | Interference Signal Cancellation Over A Broad Frequency Range |
US9894612B1 (en) | 2016-11-03 | 2018-02-13 | Corning Optical Communications Wireless Ltd | Reducing power consumption in a remote unit of a wireless distribution system (WDS) for intermodulation product suppression |
US10051643B2 (en) | 2011-08-17 | 2018-08-14 | Skyline Partners Technology Llc | Radio with interference measurement during a blanking interval |
CN108900212A (en) * | 2012-07-30 | 2018-11-27 | 光子系统股份有限公司 | The communication system sent and received simultaneously with any frequency in aperture |
US10321357B1 (en) * | 2016-07-16 | 2019-06-11 | GenXComm, Inc. | Interference cancellation methods and apparatus |
US10355771B1 (en) | 2017-05-22 | 2019-07-16 | Resonant Sciences, LLC | RF repeater and mobile unit with cancellation of interference from a repeated signal |
US10548132B2 (en) | 2011-08-17 | 2020-01-28 | Skyline Partners Technology Llc | Radio with antenna array and multiple RF bands |
US10708918B2 (en) | 2011-08-17 | 2020-07-07 | Skyline Partners Technology Llc | Electronic alignment using signature emissions for backhaul radios |
US10716111B2 (en) | 2011-08-17 | 2020-07-14 | Skyline Partners Technology Llc | Backhaul radio with adaptive beamforming and sample alignment |
US10764891B2 (en) | 2011-08-17 | 2020-09-01 | Skyline Partners Technology Llc | Backhaul radio with advanced error recovery |
US11150409B2 (en) | 2018-12-27 | 2021-10-19 | GenXComm, Inc. | Saw assisted facet etch dicing |
US11215755B2 (en) | 2019-09-19 | 2022-01-04 | GenXComm, Inc. | Low loss, polarization-independent, large bandwidth mode converter for edge coupling |
US11309965B2 (en) | 2019-07-15 | 2022-04-19 | GenXComm, Inc. | Efficiently combining multiple taps of an optical filter |
US11469821B2 (en) | 2015-12-13 | 2022-10-11 | GenXComm, Inc. | Interference cancellation methods and apparatus |
US11539394B2 (en) | 2019-10-29 | 2022-12-27 | GenXComm, Inc. | Self-interference mitigation in in-band full-duplex communication systems |
US11539392B2 (en) | 2012-07-30 | 2022-12-27 | Photonic Systems, Inc. | Same-aperture any-frequency simultaneous transmit and receive communication system |
US11796737B2 (en) | 2020-08-10 | 2023-10-24 | GenXComm, Inc. | Co-manufacturing of silicon-on-insulator waveguides and silicon nitride waveguides for hybrid photonic integrated circuits |
US11838056B2 (en) | 2021-10-25 | 2023-12-05 | GenXComm, Inc. | Hybrid photonic integrated circuits for ultra-low phase noise signal generators |
Families Citing this family (50)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US6704349B1 (en) | 2000-01-18 | 2004-03-09 | Ditrans Corporation | Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth |
US6704352B1 (en) * | 2000-05-04 | 2004-03-09 | Samsung Electronics Co., Ltd. | High accuracy receiver forward and reflected path test injection circuit |
JP4505981B2 (en) * | 2000-10-24 | 2010-07-21 | ソニー株式会社 | Spread spectrum receiver |
DE10102709B4 (en) * | 2001-01-22 | 2014-02-06 | Rohde & Schwarz Gmbh & Co. Kg | Method and apparatus for synchronization to a pilot sequence of a CDMA signal |
US7346134B2 (en) * | 2001-05-15 | 2008-03-18 | Finesse Wireless, Inc. | Radio receiver |
FR2828617B1 (en) * | 2001-08-09 | 2004-01-30 | Sagem | MOBILE TELEPHONE WITH POWER CONTROLLED ASSOCIATED METHOD |
KR100457924B1 (en) * | 2002-10-07 | 2004-11-18 | 한국전자통신연구원 | Quadrature demodulating apparatus for compensating gain and phase imbalances between in-phase and quadrature-phase components |
DE10337417B3 (en) * | 2003-08-14 | 2005-05-19 | Siemens Ag | Communication terminal for multiple reception and echo cancellation |
EP1508975A1 (en) * | 2003-08-18 | 2005-02-23 | Alcatel | Radio frequency device using circulator and echo canceller for cancelling transmission leakage signal in the reception chain |
US7711329B2 (en) | 2003-11-12 | 2010-05-04 | Qualcomm, Incorporated | Adaptive filter for transmit leakage signal rejection |
WO2005076489A1 (en) * | 2004-02-04 | 2005-08-18 | Brother Kogyo Kabushiki Kaisha | Wireless tag communication device |
WO2006023319A1 (en) * | 2004-08-24 | 2006-03-02 | Bae Systems Information And Electronic Systems | Duplexer for simultaneous transmit and receive radar systems |
US7483479B2 (en) * | 2004-09-16 | 2009-01-27 | Keyeye Communications | Scaled signal processing elements for reduced filter tap noise |
WO2006068635A1 (en) * | 2004-11-15 | 2006-06-29 | Qualcomm Incorporated | Adaptive filter for transmit leakage signal rejection |
KR100925131B1 (en) | 2004-11-15 | 2009-11-05 | 퀄컴 인코포레이티드 | Adaptive filter for transmit leakage signal rejection |
JP4214992B2 (en) * | 2004-12-13 | 2009-01-28 | パナソニック株式会社 | High frequency receiver, integrated circuit used therefor, portable device using the same, transmitter used therefor, and method for manufacturing the high frequency receiver and the portable device |
US7551694B2 (en) * | 2005-01-20 | 2009-06-23 | Marvell World Trade Ltd. | Limiter based analog demodulator |
JP4600114B2 (en) * | 2005-03-28 | 2010-12-15 | ブラザー工業株式会社 | Wireless tag communication device |
US20070082617A1 (en) * | 2005-10-11 | 2007-04-12 | Crestcom, Inc. | Transceiver with isolation-filter compensation and method therefor |
KR100653199B1 (en) * | 2005-11-18 | 2006-12-05 | 삼성전자주식회사 | Rf receiving apparatus and method for removing leakage component of received signal using local signal |
KR101365826B1 (en) * | 2006-10-17 | 2014-02-21 | 인터디지탈 테크날러지 코포레이션 | Transceiver with hybrid adaptive interference canceller for removing transmitter generated noise |
KR101452999B1 (en) * | 2008-01-25 | 2014-10-21 | 삼성전자주식회사 | Apparatus and method for calibration in multi-antenna system |
US8010055B2 (en) * | 2008-02-13 | 2011-08-30 | Viasat, Inc. | Method and apparatus for RF communication system signal to noise ratio improvement |
KR20090087629A (en) * | 2008-02-13 | 2009-08-18 | 삼성전자주식회사 | Wireless receiver and wireless communication system including the same |
US8175535B2 (en) | 2008-02-27 | 2012-05-08 | Telefonaktiebolaget Lm Ericsson (Publ) | Active cancellation of transmitter leakage in a wireless transceiver |
US20090323856A1 (en) * | 2008-06-27 | 2009-12-31 | Crestcom, Inc. | Transmit-canceling transceiver responsive to heat signal and method therefor |
EP2226946A3 (en) * | 2009-03-04 | 2013-01-02 | Technische Universität Dresden | Method for compensating for crosstalk between a transmission signal generated in a transmission branch and a reception branch |
JP5166372B2 (en) * | 2009-08-14 | 2013-03-21 | クゥアルコム・インコーポレイテッド | Adaptive filter for removing transmission leakage signal |
US8320868B2 (en) * | 2010-02-11 | 2012-11-27 | Mediatek Singapore Pte. Ltd. | Integrated circuits, communication units and methods of cancellation of intermodulation distortion |
US8471761B1 (en) | 2010-04-23 | 2013-06-25 | Akela, Inc. | Wideband radar nulling system |
KR101723233B1 (en) * | 2010-08-17 | 2017-04-04 | 엘지이노텍 주식회사 | Wireless communication apparatus improved tx/rx isolation |
US8600435B2 (en) * | 2011-04-15 | 2013-12-03 | Intel Mobile Communications GmbH | Multi-standard transceiver, device and method |
US8818299B2 (en) * | 2011-06-01 | 2014-08-26 | Andrew Llc | Broadband distributed antenna system with non-duplexer isolator sub-system |
GB2502279B (en) | 2012-05-21 | 2014-07-09 | Aceaxis Ltd | Reduction of intermodulation products |
US9209840B2 (en) | 2012-07-30 | 2015-12-08 | Photonic Systems, Inc. | Same-aperture any-frequency simultaneous transmit and receive communication system |
US10374656B2 (en) | 2012-07-30 | 2019-08-06 | Photonic Systems, Inc. | Same-aperture any-frequency simultaneous transmit and receive communication system |
US9025646B2 (en) * | 2013-03-14 | 2015-05-05 | Qualcomm, Incorporated | Transmit leakage cancellation |
CN104168234B (en) | 2013-05-16 | 2018-04-10 | 中兴通讯股份有限公司 | A kind of signal cancellation method and device of wireless telecommunication system |
US9236997B2 (en) * | 2013-09-23 | 2016-01-12 | Broadcom Corporation | Wireless transceiver with circulator-based quadrature duplexer and methods for use therewith |
US10128879B2 (en) * | 2014-03-31 | 2018-11-13 | Intel IP Corporation | Enhanced receive sensitivity for concurrent communications |
EP3407675B1 (en) * | 2014-04-15 | 2022-03-09 | CommScope Technologies LLC | Wideband remote unit for distributed antenna system |
US9693250B1 (en) | 2015-07-16 | 2017-06-27 | Viasat, Inc. | Systems and methods for monitoring electromagnetic compatibility |
US9564932B1 (en) | 2015-07-16 | 2017-02-07 | LGS Innovations LLC | Software defined radio front end |
FR3052311B1 (en) * | 2016-06-06 | 2019-08-02 | Airbus Ds Slc | DEVICE AND METHOD FOR PROCESSING A SIGNAL RECEIVED BY A PERTURBE RECEIVER BY A TRANSMITTER |
US10419062B2 (en) * | 2016-10-12 | 2019-09-17 | Massachusetts Institute Of Technology | Simultaneous transmit and receive with digital phased arrays |
US10511345B2 (en) | 2017-05-24 | 2019-12-17 | Capacicom Ltd. | Downstream interference suppression in full-duplex communications |
US11641230B2 (en) * | 2017-09-26 | 2023-05-02 | Photonic Systems, Inc. | Single-channel, full-time full-duplex wireless signal transmission system |
US10505571B1 (en) * | 2018-06-27 | 2019-12-10 | Capacicom Ltd. | Estimation of interference suppression filters using selective signal switching |
CN112543037B (en) * | 2019-09-20 | 2022-05-24 | 华为技术有限公司 | Communication equipment, radio frequency interference elimination method and device |
US11206122B1 (en) * | 2020-11-29 | 2021-12-21 | Silicon Laboratories Inc. | Variable rate sampling for AGC in a bluetooth receiver using connection state and access address field |
Citations (28)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3500000A (en) * | 1966-10-31 | 1970-03-10 | Myldred P Kelly | Self-adaptive echo canceller |
US4475243A (en) * | 1982-12-21 | 1984-10-02 | Motorola, Inc. | Isolation method and apparatus for a same frequency repeater |
US4535476A (en) * | 1982-12-01 | 1985-08-13 | At&T Bell Laboratories | Offset geometry, interference canceling receiver |
US4720712A (en) * | 1985-08-12 | 1988-01-19 | Raytheon Company | Adaptive beam forming apparatus |
US4953182A (en) * | 1987-09-03 | 1990-08-28 | U.S. Philips Corporation | Gain and phase correction in a dual branch receiver |
US5249203A (en) * | 1991-02-25 | 1993-09-28 | Rockwell International Corporation | Phase and gain error control system for use in an i/q direct conversion receiver |
US5396517A (en) * | 1993-03-04 | 1995-03-07 | Adtran | Transversal filter useable in echo canceler, decision feedback equalizer applications for minimizing non-linear distortion in signals conveyed over full duplex two-wire communication link |
US5422889A (en) * | 1992-10-28 | 1995-06-06 | Alcatel N.V. | Offset correction circuit |
US5557642A (en) * | 1992-08-25 | 1996-09-17 | Wireless Access, Inc. | Direct conversion receiver for multiple protocols |
US5596439A (en) * | 1995-08-01 | 1997-01-21 | Viasat, Inc. | Self-interference cancellation for two-party relayed communication |
US5604929A (en) * | 1995-04-21 | 1997-02-18 | Rockwell International | System for correcting quadrature gain and phase errors in a direct conversion single sideband receiver independent of the character of the modulated signal |
US5675613A (en) * | 1995-02-02 | 1997-10-07 | Nippon Telegraph & Telephone Corporation | Distortion compensator |
US5691978A (en) * | 1995-04-07 | 1997-11-25 | Signal Science, Inc. | Self-cancelling full-duplex RF communication system |
US5705949A (en) * | 1996-09-13 | 1998-01-06 | U.S. Robotics Access Corp. | Compensation method for I/Q channel imbalance errors |
US5793801A (en) * | 1996-07-09 | 1998-08-11 | Telefonaktiebolaget Lm Ericsson | Frequency domain signal reconstruction compensating for phase adjustments to a sampling signal |
US5828955A (en) * | 1995-08-30 | 1998-10-27 | Rockwell Semiconductor Systems, Inc. | Near direct conversion receiver and method for equalizing amplitude and phase therein |
US5847619A (en) * | 1996-03-14 | 1998-12-08 | Nec Corporation | Method and system for calibrating a quadrature phase modulator |
US5861837A (en) * | 1997-03-19 | 1999-01-19 | Northrop Grumman Corporation | Poly-frequency CW doppler radar system with leakage cancellation and method |
US5872540A (en) * | 1997-06-26 | 1999-02-16 | Electro-Radiation Incorporated | Digital interference suppression system for radio frequency interference cancellation |
US5926135A (en) * | 1998-01-08 | 1999-07-20 | Lucent Technologies | Steerable nulling of wideband interference signals |
US5952965A (en) * | 1998-07-21 | 1999-09-14 | Marconi Aerospace Systems Inc. Advanced Systems Division | Adaptive main beam nulling using array antenna auxiliary patterns |
US5999800A (en) * | 1996-04-18 | 1999-12-07 | Korea Telecom Freetel Co., Ltd. | Design technique of an array antenna, and telecommunication system and method utilizing the array antenna |
US6169912B1 (en) * | 1999-03-31 | 2001-01-02 | Pericom Semiconductor Corp. | RF front-end with signal cancellation using receiver signal to eliminate duplexer for a cordless phone |
US6240128B1 (en) * | 1998-06-11 | 2001-05-29 | Agere Systems Guardian Corp. | Enhanced echo canceler |
US6289048B1 (en) * | 2000-01-06 | 2001-09-11 | Cubic Communications, Inc. | Apparatus and method for improving dynamic range in a receiver |
US6340883B1 (en) * | 1998-09-03 | 2002-01-22 | Sony/Tektronik Corporation | Wide band IQ splitting apparatus and calibration method therefor with balanced amplitude and phase between I and Q |
US6421377B1 (en) * | 1998-05-13 | 2002-07-16 | Globespanvirata, Inc. | System and method for echo cancellation over asymmetric spectra |
US6714584B1 (en) * | 1998-04-07 | 2004-03-30 | Nec Corporation | CDMA adaptive antenna receiving apparatus and communication system |
Family Cites Families (1)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US6704349B1 (en) | 2000-01-18 | 2004-03-09 | Ditrans Corporation | Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth |
-
2000
- 2000-01-18 US US09/487,396 patent/US6704349B1/en not_active Expired - Lifetime
-
2001
- 2001-01-18 WO PCT/US2001/001896 patent/WO2001054290A2/en active Application Filing
- 2001-01-18 JP JP2001553672A patent/JP2003520549A/en active Pending
- 2001-01-18 EP EP01903160A patent/EP1249077A2/en not_active Withdrawn
- 2001-01-18 AU AU2001231011A patent/AU2001231011A1/en not_active Abandoned
-
2004
- 2004-01-15 US US10/758,355 patent/US20040151238A1/en not_active Abandoned
Patent Citations (28)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3500000A (en) * | 1966-10-31 | 1970-03-10 | Myldred P Kelly | Self-adaptive echo canceller |
US4535476A (en) * | 1982-12-01 | 1985-08-13 | At&T Bell Laboratories | Offset geometry, interference canceling receiver |
US4475243A (en) * | 1982-12-21 | 1984-10-02 | Motorola, Inc. | Isolation method and apparatus for a same frequency repeater |
US4720712A (en) * | 1985-08-12 | 1988-01-19 | Raytheon Company | Adaptive beam forming apparatus |
US4953182A (en) * | 1987-09-03 | 1990-08-28 | U.S. Philips Corporation | Gain and phase correction in a dual branch receiver |
US5249203A (en) * | 1991-02-25 | 1993-09-28 | Rockwell International Corporation | Phase and gain error control system for use in an i/q direct conversion receiver |
US5557642A (en) * | 1992-08-25 | 1996-09-17 | Wireless Access, Inc. | Direct conversion receiver for multiple protocols |
US5422889A (en) * | 1992-10-28 | 1995-06-06 | Alcatel N.V. | Offset correction circuit |
US5396517A (en) * | 1993-03-04 | 1995-03-07 | Adtran | Transversal filter useable in echo canceler, decision feedback equalizer applications for minimizing non-linear distortion in signals conveyed over full duplex two-wire communication link |
US5675613A (en) * | 1995-02-02 | 1997-10-07 | Nippon Telegraph & Telephone Corporation | Distortion compensator |
US5691978A (en) * | 1995-04-07 | 1997-11-25 | Signal Science, Inc. | Self-cancelling full-duplex RF communication system |
US5604929A (en) * | 1995-04-21 | 1997-02-18 | Rockwell International | System for correcting quadrature gain and phase errors in a direct conversion single sideband receiver independent of the character of the modulated signal |
US5596439A (en) * | 1995-08-01 | 1997-01-21 | Viasat, Inc. | Self-interference cancellation for two-party relayed communication |
US5828955A (en) * | 1995-08-30 | 1998-10-27 | Rockwell Semiconductor Systems, Inc. | Near direct conversion receiver and method for equalizing amplitude and phase therein |
US5847619A (en) * | 1996-03-14 | 1998-12-08 | Nec Corporation | Method and system for calibrating a quadrature phase modulator |
US5999800A (en) * | 1996-04-18 | 1999-12-07 | Korea Telecom Freetel Co., Ltd. | Design technique of an array antenna, and telecommunication system and method utilizing the array antenna |
US5793801A (en) * | 1996-07-09 | 1998-08-11 | Telefonaktiebolaget Lm Ericsson | Frequency domain signal reconstruction compensating for phase adjustments to a sampling signal |
US5705949A (en) * | 1996-09-13 | 1998-01-06 | U.S. Robotics Access Corp. | Compensation method for I/Q channel imbalance errors |
US5861837A (en) * | 1997-03-19 | 1999-01-19 | Northrop Grumman Corporation | Poly-frequency CW doppler radar system with leakage cancellation and method |
US5872540A (en) * | 1997-06-26 | 1999-02-16 | Electro-Radiation Incorporated | Digital interference suppression system for radio frequency interference cancellation |
US5926135A (en) * | 1998-01-08 | 1999-07-20 | Lucent Technologies | Steerable nulling of wideband interference signals |
US6714584B1 (en) * | 1998-04-07 | 2004-03-30 | Nec Corporation | CDMA adaptive antenna receiving apparatus and communication system |
US6421377B1 (en) * | 1998-05-13 | 2002-07-16 | Globespanvirata, Inc. | System and method for echo cancellation over asymmetric spectra |
US6240128B1 (en) * | 1998-06-11 | 2001-05-29 | Agere Systems Guardian Corp. | Enhanced echo canceler |
US5952965A (en) * | 1998-07-21 | 1999-09-14 | Marconi Aerospace Systems Inc. Advanced Systems Division | Adaptive main beam nulling using array antenna auxiliary patterns |
US6340883B1 (en) * | 1998-09-03 | 2002-01-22 | Sony/Tektronik Corporation | Wide band IQ splitting apparatus and calibration method therefor with balanced amplitude and phase between I and Q |
US6169912B1 (en) * | 1999-03-31 | 2001-01-02 | Pericom Semiconductor Corp. | RF front-end with signal cancellation using receiver signal to eliminate duplexer for a cordless phone |
US6289048B1 (en) * | 2000-01-06 | 2001-09-11 | Cubic Communications, Inc. | Apparatus and method for improving dynamic range in a receiver |
Cited By (139)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20050069394A1 (en) * | 2003-09-26 | 2005-03-31 | Dyer Kenneth C. | Method, device and system for output impedance calibration that invariably maximizes hybrid performance |
US7031456B2 (en) | 2003-09-26 | 2006-04-18 | Key Eye | Method, device and system for output impedance calibration that invariably maximizes hybrid performance |
US7738534B2 (en) | 2003-11-24 | 2010-06-15 | Telefonaktiebolaget Lm Ericsson (Publ) | Multi-transmitter interference suppression using code-specific combining |
US20070104254A1 (en) * | 2003-11-24 | 2007-05-10 | Bottomley Gregory E | Multi-Transmitter Interference Suppression Using Code-Specific Combining |
US20050135461A1 (en) * | 2003-12-22 | 2005-06-23 | Nokia Corporation | Deconvolution searcher for wireless communication system |
US7756191B2 (en) * | 2003-12-22 | 2010-07-13 | Nokia Corporation | Deconvolution searcher for wireless communication system |
US7406300B2 (en) * | 2004-07-29 | 2008-07-29 | Lucent Technologies Inc. | Extending wireless communication RF coverage inside building |
US20060025072A1 (en) * | 2004-07-29 | 2006-02-02 | Lucent Technologies, Inc. | Extending wireless communication RF coverage inside building |
US20070015468A1 (en) * | 2005-04-14 | 2007-01-18 | Kouki Ammar B | Balanced active and passive duplexers |
US8364092B2 (en) * | 2005-04-14 | 2013-01-29 | Ecole De Technologie Superieure | Balanced active and passive duplexers |
US20070280385A1 (en) * | 2006-05-30 | 2007-12-06 | Benq Corporation | Method and apparatus of receiving signals and wireless multimode wideband receiver |
US20090016545A1 (en) * | 2006-08-23 | 2009-01-15 | Quellan, Inc. | Pre-configuration and control of radio frequency noise cancellation |
US8315583B2 (en) | 2006-08-23 | 2012-11-20 | Quellan, Inc. | Pre-configuration and control of radio frequency noise cancellation |
US20080062029A1 (en) * | 2006-09-12 | 2008-03-13 | Xu Changqing | Apparatus and method for demodulating a modulated signal |
US7541966B2 (en) * | 2006-09-12 | 2009-06-02 | Oki Techno Centre (Singapore) Pte Ltd | Apparatus and method for demodulating a modulated signal |
US20080130719A1 (en) * | 2006-12-05 | 2008-06-05 | Bottomley Gregory E | Method and Apparatus for Suppressing Interference Based on Channelization Code Power Estimation with Bias Removal |
US7751463B2 (en) | 2006-12-05 | 2010-07-06 | Telefonaktiebolaget Lm Ericsson (Publ) | Method and apparatus for suppressing interference based on channelization code power estimation with bias removal |
US8805298B2 (en) * | 2007-01-30 | 2014-08-12 | Crestcom, Inc. | Transceiver with compensation for transmit signal leakage and method therefor |
US20100048146A1 (en) * | 2007-01-30 | 2010-02-25 | Crestcom, Inc. | Transceiver with Compensation for Transmit Signal Leakage and Method Therefor |
US20090115579A1 (en) * | 2007-11-06 | 2009-05-07 | Microelectronics Technology Inc. | Signal processing apparatus for receiving rfid signal and method thereof |
US20090116455A1 (en) * | 2007-11-06 | 2009-05-07 | Bottomley Gregory E | Method and Apparatus for Code Power Parameter Estimation for Received Signal Processing |
US7995641B2 (en) | 2007-11-06 | 2011-08-09 | Telefonaktiebolaget Lm Ericsson (Publ) | Method and apparatus for code power parameter estimation for received signal processing |
WO2010011623A1 (en) * | 2008-07-21 | 2010-01-28 | Quellan, Inc. | Pre-configuration and control of radio frequency noise cancellation |
US20110228828A1 (en) * | 2008-11-21 | 2011-09-22 | Wei Wang | Method and device for cancelling transmitter interference in transceiver, and transceiver |
US8331509B2 (en) | 2008-11-21 | 2012-12-11 | Huawei Technologies Co., Ltd. | Method and device for cancelling transmitter interference in transceiver, and transceiver |
CN101420246B (en) * | 2008-11-21 | 2013-09-11 | 华为技术有限公司 | Method, apparatus and transceiver for counteracting transmission interference by the transceiver |
US8199681B2 (en) * | 2008-12-12 | 2012-06-12 | General Electric Company | Software radio frequency canceller |
US20100150032A1 (en) * | 2008-12-12 | 2010-06-17 | General Electric Company | Software radio frequency canceller |
US9178559B2 (en) | 2008-12-12 | 2015-11-03 | St Ericsson Sa | Method and system of calibration of a second order intermodulation intercept point of a radio transceiver |
US20100150033A1 (en) * | 2008-12-16 | 2010-06-17 | General Electric Company | Software radio frequency canceller |
US9130747B2 (en) * | 2008-12-16 | 2015-09-08 | General Electric Company | Software radio frequency canceller |
CN102771054A (en) * | 2010-02-11 | 2012-11-07 | 联发科技(新加坡)私人有限公司 | Integrated circuits, communication units and methods of cancellation of intermodulation distortion |
US20110243202A1 (en) * | 2010-04-01 | 2011-10-06 | Ismail Lakkis | Cancellation System for Millimeter-Wave Radar |
US8611401B2 (en) * | 2010-04-01 | 2013-12-17 | Adeptence, Llc | Cancellation system for millimeter-wave radar |
US20150036773A1 (en) * | 2010-04-01 | 2015-02-05 | Ismail Lakkis | Cancellation System for Millimeter-Wave Radar |
US8928542B2 (en) | 2011-08-17 | 2015-01-06 | CBF Networks, Inc. | Backhaul radio with an aperture-fed antenna assembly |
US9282560B2 (en) | 2011-08-17 | 2016-03-08 | CBF Networks, Inc. | Full duplex backhaul radio with transmit beamforming and SC-FDE modulation |
US11160078B2 (en) | 2011-08-17 | 2021-10-26 | Skyline Partners Technology, Llc | Backhaul radio with adaptive beamforming and sample alignment |
US9713155B2 (en) | 2011-08-17 | 2017-07-18 | CBF Networks, Inc. | Radio with antenna array and multiple RF bands |
US11271613B2 (en) | 2011-08-17 | 2022-03-08 | Skyline Partners Technology Llc | Radio with spatially-offset directional antenna sub-arrays |
US8811365B2 (en) | 2011-08-17 | 2014-08-19 | CBF Networks, Inc. | Intelligent backhaul radio |
US8824442B2 (en) | 2011-08-17 | 2014-09-02 | CBF Networks, Inc. | Intelligent backhaul radio with adaptive channel bandwidth control |
US11134491B2 (en) | 2011-08-17 | 2021-09-28 | Skyline Partners Technology Llc | Radio with antenna array and multiple RF bands |
US9713019B2 (en) | 2011-08-17 | 2017-07-18 | CBF Networks, Inc. | Self organizing backhaul radio |
US8872715B2 (en) | 2011-08-17 | 2014-10-28 | CBF Networks, Inc. | Backhaul radio with a substrate tab-fed antenna assembly |
US10051643B2 (en) | 2011-08-17 | 2018-08-14 | Skyline Partners Technology Llc | Radio with interference measurement during a blanking interval |
US9712216B2 (en) | 2011-08-17 | 2017-07-18 | CBF Networks, Inc. | Radio with spatially-offset directional antenna sub-arrays |
US10764891B2 (en) | 2011-08-17 | 2020-09-01 | Skyline Partners Technology Llc | Backhaul radio with advanced error recovery |
US11283192B2 (en) | 2011-08-17 | 2022-03-22 | Skyline Partners Technology Llc | Aperture-fed, stacked-patch antenna assembly |
US8982772B2 (en) | 2011-08-17 | 2015-03-17 | CBF Networks, Inc. | Radio transceiver with improved radar detection |
US9713157B2 (en) | 2011-08-17 | 2017-07-18 | CBF Networks, Inc. | Method for installing a backhaul link with alignment signals |
US9001809B2 (en) | 2011-08-17 | 2015-04-07 | CBF Networks, Inc. | Intelligent backhaul radio with transmit and receive antenna arrays |
US10735979B2 (en) | 2011-08-17 | 2020-08-04 | Skyline Partners Technology Llc | Self organizing backhaul radio |
US9049611B2 (en) | 2011-08-17 | 2015-06-02 | CBF Networks, Inc. | Backhaul radio with extreme interference protection |
US9055463B2 (en) | 2011-08-17 | 2015-06-09 | CBF Networks, Inc. | Intelligent backhaul radio with receiver performance enhancement |
US10135501B2 (en) | 2011-08-17 | 2018-11-20 | Skyline Partners Technology Llc | Radio with spatially-offset directional antenna sub-arrays |
US10720969B2 (en) | 2011-08-17 | 2020-07-21 | Skyline Partners Technology Llc | Radio with spatially-offset directional antenna sub-arrays |
US9655133B2 (en) | 2011-08-17 | 2017-05-16 | CBF Networks, Inc. | Radio with interference measurement during a blanking interval |
US9178558B2 (en) | 2011-08-17 | 2015-11-03 | CBF Networks, Inc. | Backhaul radio with horizontally or vertically arranged receive antenna arrays |
US8467363B2 (en) | 2011-08-17 | 2013-06-18 | CBF Networks, Inc. | Intelligent backhaul radio and antenna system |
US10716111B2 (en) | 2011-08-17 | 2020-07-14 | Skyline Partners Technology Llc | Backhaul radio with adaptive beamforming and sample alignment |
US9609530B2 (en) | 2011-08-17 | 2017-03-28 | CBF Networks, Inc. | Aperture-fed, stacked-patch antenna assembly |
US10708918B2 (en) | 2011-08-17 | 2020-07-07 | Skyline Partners Technology Llc | Electronic alignment using signature emissions for backhaul radios |
US11166280B2 (en) | 2011-08-17 | 2021-11-02 | Skyline Partners Technology, Llc | Backhaul radio with advanced error recovery |
US9313674B2 (en) | 2011-08-17 | 2016-04-12 | CBF Networks, Inc. | Backhaul radio with extreme interference protection |
US10237760B2 (en) | 2011-08-17 | 2019-03-19 | Skyline Partners Technology Llc | Self organizing backhaul radio |
US9345036B2 (en) | 2011-08-17 | 2016-05-17 | CBF Networks, Inc. | Full duplex radio transceiver with remote radar detection |
US9350411B2 (en) | 2011-08-17 | 2016-05-24 | CBF Networks, Inc. | Full duplex backhaul radio with MIMO antenna array |
US11343684B2 (en) | 2011-08-17 | 2022-05-24 | Skyline Partners Technology Llc | Self organizing backhaul radio |
US10548132B2 (en) | 2011-08-17 | 2020-01-28 | Skyline Partners Technology Llc | Radio with antenna array and multiple RF bands |
US9408215B2 (en) | 2011-08-17 | 2016-08-02 | CBF Networks, Inc. | Full duplex backhaul radio with transmit beamforming |
US10506611B2 (en) | 2011-08-17 | 2019-12-10 | Skyline Partners Technology Llc | Radio with interference measurement during a blanking interval |
US9474080B2 (en) | 2011-08-17 | 2016-10-18 | CBF Networks, Inc. | Full duplex backhaul radio with interference measurement during a blanking interval |
US10313898B2 (en) | 2011-08-17 | 2019-06-04 | Skyline Partners Technology Llc | Aperture-fed, stacked-patch antenna assembly |
US9572163B2 (en) | 2011-08-17 | 2017-02-14 | CBF Networks, Inc. | Hybrid band radio with adaptive antenna arrays |
US9577733B2 (en) | 2011-08-17 | 2017-02-21 | CBF Networks, Inc. | Method for installing a backhaul link with multiple antenna patterns |
US9577700B2 (en) | 2011-08-17 | 2017-02-21 | CBF Networks, Inc. | Radio with asymmetrical directional antenna sub-arrays |
US9578643B2 (en) | 2011-08-17 | 2017-02-21 | CBF Networks, Inc. | Backhaul radio with antenna array and multiple RF carrier frequencies |
US10306635B2 (en) | 2011-08-17 | 2019-05-28 | Skyline Partners Technology Llc | Hybrid band radio with multiple antenna arrays |
US9226315B2 (en) | 2011-10-11 | 2015-12-29 | CBF Networks, Inc. | Intelligent backhaul radio with multi-interface switching |
US10785754B2 (en) | 2011-10-11 | 2020-09-22 | Skyline Partners Technology Llc | Method for deploying a backhaul radio with antenna array |
US8830943B2 (en) | 2011-10-11 | 2014-09-09 | CBF Networks, Inc. | Intelligent backhaul management system |
US8761100B2 (en) | 2011-10-11 | 2014-06-24 | CBF Networks, Inc. | Intelligent backhaul system |
US9325398B2 (en) | 2012-02-10 | 2016-04-26 | CBF Networks, Inc. | Method for installing a backhaul radio with an antenna array |
US9179240B2 (en) | 2012-02-10 | 2015-11-03 | CBF Networks, Inc. | Transmit co-channel spectrum sharing |
US8502733B1 (en) | 2012-02-10 | 2013-08-06 | CBF Networks, Inc. | Transmit co-channel spectrum sharing |
US10736110B2 (en) | 2012-02-10 | 2020-08-04 | Skyline Partners Technology Llc | Method for installing a fixed wireless access link with alignment signals |
US10129888B2 (en) | 2012-02-10 | 2018-11-13 | Skyline Partners Technology Llc | Method for installing a fixed wireless access link with alignment signals |
US9374822B2 (en) | 2012-04-16 | 2016-06-21 | CBF Networks, Inc. | Method for installing a hybrid band radio |
US9226295B2 (en) | 2012-04-16 | 2015-12-29 | CBF Networks, Inc. | Hybrid band radio with data direction determined by a link performance metric |
US10932267B2 (en) | 2012-04-16 | 2021-02-23 | Skyline Partners Technology Llc | Hybrid band radio with multiple antenna arrays |
US8942216B2 (en) | 2012-04-16 | 2015-01-27 | CBF Networks, Inc. | Hybrid band intelligent backhaul radio |
US20150103745A1 (en) * | 2012-06-21 | 2015-04-16 | CBF Networks, Inc. | Intelligent backhaul radio with co-band zero division duplexing |
US20140106688A1 (en) * | 2012-06-21 | 2014-04-17 | CBF Networks, Inc. | Intelligent backhaul radio with zero division duplexing |
US11343060B2 (en) * | 2012-06-21 | 2022-05-24 | Skyline Partners Technology Llc | Zero division duplexing mimo radio with adaptable RF and/or baseband cancellation |
US8422540B1 (en) * | 2012-06-21 | 2013-04-16 | CBF Networks, Inc. | Intelligent backhaul radio with zero division duplexing |
US8638839B2 (en) * | 2012-06-21 | 2014-01-28 | CBF Networks, Inc. | Intelligent backhaul radio with co-band zero division duplexing |
US10063363B2 (en) * | 2012-06-21 | 2018-08-28 | Skyline Partners Technology Llc | Zero division duplexing MIMO radio with adaptable RF and/or baseband cancellation |
US8948235B2 (en) * | 2012-06-21 | 2015-02-03 | CBF Networks, Inc. | Intelligent backhaul radio with co-band zero division duplexing utilizing transmitter to receiver antenna isolation adaptation |
US20220286266A1 (en) * | 2012-06-21 | 2022-09-08 | Skyline Partners Technology Llc | Zero division duplexing mimo radio with adaptable rf and/or baseband cancellation |
US9490918B2 (en) * | 2012-06-21 | 2016-11-08 | CBF Networks, Inc. | Zero division duplexing MIMO backhaul radio with adaptable RF and/or baseband cancellation |
US20170054545A1 (en) * | 2012-06-21 | 2017-02-23 | CBF Networks, Inc. | Zero division duplexing mimo radio with adaptable rf and/or baseband cancellation |
US11539392B2 (en) | 2012-07-30 | 2022-12-27 | Photonic Systems, Inc. | Same-aperture any-frequency simultaneous transmit and receive communication system |
CN108900212A (en) * | 2012-07-30 | 2018-11-27 | 光子系统股份有限公司 | The communication system sent and received simultaneously with any frequency in aperture |
US10651886B2 (en) | 2012-07-30 | 2020-05-12 | Photonic Systems, Inc. | Same-aperture any-frequency simultaneous transmit and receive communication system |
US10879950B2 (en) | 2012-07-30 | 2020-12-29 | Photonic Systems, Inc. | Same-aperture any-frequency simultaneous transmit and receive communication system |
USD704174S1 (en) | 2012-08-14 | 2014-05-06 | CBF Networks, Inc. | Intelligent backhaul radio with symmetric wing radome |
US20140161005A1 (en) * | 2012-11-13 | 2014-06-12 | Stephane Laurent-Michel | Method and system for signal dynamic range improvement for frequency-division duplex communication systems |
US9391717B2 (en) * | 2012-11-13 | 2016-07-12 | Stephane Laurent-Michel | Method and system for signal dynamic range improvement for frequency-division duplex communication systems |
US9252831B2 (en) * | 2013-03-01 | 2016-02-02 | Qualcomm Incorporated | Multi-tap adaptive filter for transmit signal leakage cancellation |
US20140247757A1 (en) * | 2013-03-01 | 2014-09-04 | Qualcomm Incorporated | Multi-tap adaptive filter for transmit signal leakage cancellation |
US9876530B2 (en) | 2013-12-05 | 2018-01-23 | Skyline Partners Technology, Llc | Advanced backhaul services |
US10284253B2 (en) | 2013-12-05 | 2019-05-07 | Skyline Partners Technology Llc | Advanced backhaul services |
US10700733B2 (en) | 2013-12-05 | 2020-06-30 | Skyline Partners Technology Llc | Advanced backhaul services |
US11303322B2 (en) | 2013-12-05 | 2022-04-12 | Skyline Partners Technology Llc | Advanced backhaul services |
US8989762B1 (en) | 2013-12-05 | 2015-03-24 | CBF Networks, Inc. | Advanced backhaul services |
CN104967461A (en) * | 2014-03-31 | 2015-10-07 | 英特尔Ip公司 | Self-interference rejection based on correlation |
US9781612B2 (en) | 2014-03-31 | 2017-10-03 | Intel IP Corporation | Correlation-based self-interference suppression |
TWI551089B (en) * | 2014-03-31 | 2016-09-21 | 英特爾智財公司 | Correlation-based self-interference suppression |
US9603155B2 (en) * | 2015-07-31 | 2017-03-21 | Corning Optical Communications Wireless Ltd | Reducing leaked downlink interference signals in a remote unit uplink path(s) in a distributed antenna system (DAS) |
US9960850B2 (en) * | 2015-07-31 | 2018-05-01 | Corning Optical Communications Wireless Ltd | Reducing leaked downlink interference signals in a remote unit uplink path(s) in a distributed antenna system (DAS) |
US20170149504A1 (en) * | 2015-07-31 | 2017-05-25 | Corning Optical Communications Wireless Ltd | Reducing leaked downlink interference signals in a remote unit uplink path(s) in a distributed antenna system (das) |
US11469821B2 (en) | 2015-12-13 | 2022-10-11 | GenXComm, Inc. | Interference cancellation methods and apparatus |
US10084496B2 (en) * | 2016-02-23 | 2018-09-25 | Resonant Sciences, LLC | Interference signal cancellation over a broad frequency range |
US20170264327A1 (en) * | 2016-02-23 | 2017-09-14 | Resonant Sciences, LLC | Interference Signal Cancellation Over A Broad Frequency Range |
US20190253922A1 (en) * | 2016-07-16 | 2019-08-15 | GenXComm, Inc. | Interference cancellation methods and apparatus |
US10321357B1 (en) * | 2016-07-16 | 2019-06-11 | GenXComm, Inc. | Interference cancellation methods and apparatus |
US11330464B2 (en) * | 2016-07-16 | 2022-05-10 | GenXComm, Inc. | Interference cancellation methods and apparatus |
US10873877B2 (en) * | 2016-07-16 | 2020-12-22 | GenXComm, Inc. | Interference cancellation methods and apparatus |
US9894612B1 (en) | 2016-11-03 | 2018-02-13 | Corning Optical Communications Wireless Ltd | Reducing power consumption in a remote unit of a wireless distribution system (WDS) for intermodulation product suppression |
US10219220B2 (en) | 2016-11-03 | 2019-02-26 | Corning Optical Communications Wireless Ltd | Reducing power consumption in a remote unit of a wireless distribution system (WDS) for intermodulation product suppression |
US10355771B1 (en) | 2017-05-22 | 2019-07-16 | Resonant Sciences, LLC | RF repeater and mobile unit with cancellation of interference from a repeated signal |
US10587331B1 (en) | 2017-05-22 | 2020-03-10 | Resonant Sciences, LLC | RF repeater and mobile unit with cancellation of interference from a repeated signal |
US11150409B2 (en) | 2018-12-27 | 2021-10-19 | GenXComm, Inc. | Saw assisted facet etch dicing |
US11309965B2 (en) | 2019-07-15 | 2022-04-19 | GenXComm, Inc. | Efficiently combining multiple taps of an optical filter |
US11215755B2 (en) | 2019-09-19 | 2022-01-04 | GenXComm, Inc. | Low loss, polarization-independent, large bandwidth mode converter for edge coupling |
US11539394B2 (en) | 2019-10-29 | 2022-12-27 | GenXComm, Inc. | Self-interference mitigation in in-band full-duplex communication systems |
US11796737B2 (en) | 2020-08-10 | 2023-10-24 | GenXComm, Inc. | Co-manufacturing of silicon-on-insulator waveguides and silicon nitride waveguides for hybrid photonic integrated circuits |
US11838056B2 (en) | 2021-10-25 | 2023-12-05 | GenXComm, Inc. | Hybrid photonic integrated circuits for ultra-low phase noise signal generators |
Also Published As
Publication number | Publication date |
---|---|
WO2001054290A2 (en) | 2001-07-26 |
EP1249077A2 (en) | 2002-10-16 |
JP2003520549A (en) | 2003-07-02 |
WO2001054290A3 (en) | 2002-04-11 |
US6704349B1 (en) | 2004-03-09 |
AU2001231011A1 (en) | 2001-07-31 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
US6704349B1 (en) | Method and apparatus for canceling a transmit signal spectrum in a receiver bandwidth | |
US9722647B2 (en) | Calibration techniques for sigma delta transceivers | |
US11451363B2 (en) | Digital-centric full-duplex architecture | |
US6748025B1 (en) | Direct conversion delta-sigma receiver | |
TWI410060B (en) | Rejection of transmit signal leakage in wireless communication device | |
US9985809B2 (en) | Dynamic range of wideband RF front end using delta sigma converters with envelope tracking and injected digitally equalized transmit signal | |
US20050119025A1 (en) | Serial digital interface for wireless network radios and baseband integrated circuits | |
US9698845B2 (en) | High oversampling ratio dynamic element matching scheme for high dynamic range digital to RF data conversion for radio communication systems | |
US20080219331A1 (en) | Methods and apparatus for reducing the effects of DAC images in radio frequency transceivers | |
US9622181B2 (en) | Power efficient, variable sampling rate delta-sigma data converters for flexible radio communication systems | |
US9853843B2 (en) | Software programmable, multi-segment capture bandwidth, delta-sigma modulators for flexible radio communication systems | |
Katanbaf et al. | Two-Way Traffic Ahead: RF\/Analog Self-Interference Cancellation Techniques and the Challenges for Future Integrated Full-Duplex Transceivers | |
US9660690B2 (en) | Optimized data converter design using mixed semiconductor technology for flexible radio communication systems | |
WO2004098081A1 (en) | Front end of a multi-standard two-channel direct-conversion quadrature receiver | |
Kanumalli et al. | Digitally-intensive transceivers for future mobile communications—emerging trends and challenges | |
US11251880B2 (en) | CA power measurement | |
US7020221B2 (en) | System and method for an IF-sampling transceiver | |
JP5202035B2 (en) | Semiconductor integrated circuit and operation method thereof | |
Friedman et al. | Coexistence performance enhancement techniques for DRP based WLAN/WPAN and cellular radios collocated in a mobile device | |
Marković et al. | Multi-GHz Radio DSP | |
WO2000079706A1 (en) | Direct conversion delta-sigma receiver | |
KR20050116304A (en) | Direct conversion delta-sigma receiver |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
AS | Assignment |
Owner name: DITRANS IP, INC., CALIFORNIA Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:DITRANS CORPORATION;REEL/FRAME:015279/0288 Effective date: 20040413 |
|
STCB | Information on status: application discontinuation |
Free format text: ABANDONED -- FAILURE TO RESPOND TO AN OFFICE ACTION |