CN1409904A - 增强处理弱扩展频谱信号能力的强信号消除方法 - Google Patents
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Abstract
接收包含强信号和弱信号的一个CDMA编码的扩展频谱无线信号,并且计算该强信号与该弱信号的干扰以便增强跟踪该弱信号的能力。该编码调制的两个信号都是已知信号,并且能预测该弱信号。强信号的干扰被计算为该强信号的幅度和该强信号与该弱信号的预测互相关联性的乘积。该强信号可被测量、预测或通过测量和预测的两个方法组合获取。可以针对弱信号值的一个范围以及按照产生最大接收功率的预测而选择的弱信号来预测该互相关联性。
Description
发明领域
本发明涉及多址扩展频谱无线接收机,尤其涉及在出现相对较强信号的情况下具有获取和跟踪相对弱的信号的增强能力的接收机。
信号强度中的差异往往归因于信号源和接收机的相对距离,因此常把出现较近、较强信号时对跟踪弱信号的困难称之为扩展频谱多址的‘近-远’问题。当一个信号源与接收机隐蔽而另一信号源具有直达线路的位置时也将会出现此问题。一个实例是在一个建筑物内操作一个接收机,或许靠近窗口或门,由此以正常信号强度接收某些信号而其它正常信号则由该建筑物的结构所衰减。
已有技术的描述
全球定位系统(GPS)是由美国空军操作的一个无线电导航系统,用于提供为军事以及民用用户提供精确全球定位信息的双重用途。为此目的,GPS提供两种业务:精确定位业务(PPS),主要用于美国军队并且要求配备了适当PPS设备的接收机:以及普通定位业务(SPS),不如PPS精确,但可供所有用户所用,不论其是否可以使用PPS设备。美国的国防部有能力通过被称为‘选择性利用’(S/A)的算法来降低该SPS的精确度,并且采用官方定位标准,即所有这种S/A引发的误差将限制到100米水平定位误差范围(2d-RMS)。相对照,PPS是精确到22米之内。
GPS实际上包括至少24颗卫星,在围绕地球的高度近似为20000Km的六个轨道之一中。每一轨道由至少四颗卫星占用。每一GPS卫星传播能够由正确配备的GPS接收机接收的唯一无线电航向信号。该信号包含标识该具体发射卫星和导航数据的信息,例如时间和卫星位置的信息。根据一个基础电平,所有的GPS接收机通过跟踪多颗GPS卫星的距离范围而操作,并且依据纬度、经度、高度或其它等效空间座标系而确定用户的位置。
由每一卫星传播的航向信号包括两个信号:以1575.42MHz载频传播的基本链路1(L1)信号和以1227.6MHz的载频传播的辅助链路2(L2)信号。L1和L2载波信号都是由数字信号或码调制的扩展频谱信号,它″扩展″在一个规定带宽上的每一个载波信号的频谱。该L1信号由三个双相位(即±1)数字信号调制:清晰或粗略采集(C/A)码,该码是以1.023MHz的比特率(或涉及噪音码的每一个脉冲的码片)广播的短伪随机噪音(PRN)码(并且因此通过实际上把该原始信号中的每一个比特分解成1023个单独的比特或码片,在1.023MHz带宽上扩展该L1载波信号,被称为定向序列扩展频谱),并且因此以每一1毫秒重复一个码片;精确(P)码,是每周重复的长得多的PRN码,并且以C/A码的十倍码片速率传播(10.23MHz);以及一个50Hz的导引数据码(D)。C/A码总是以清晰(即未被加密)的状态传播,P码由加密(E)码加密以便形成被称为Y码的加密码。低数据速率导引码D包括用于卫星的由S/A修正的轨道参数和时钟校正信息。
当前该SPS只依据L1信号预测,但是未来的SPS信号将可用在L1和L2。当前的L1信号包含由PED调制的同相成份(其中表示逻辑XOR操作)和由C/AD调制的正交成分,并且能够针对每一卫星i表示为: 其中A表示信号功率,ω表示载波频率,而表示小的相位噪声和振荡器频率漂移(即时钟误差)成分。
广播卫星导引数据信息D和处理该数据信息D的算法在公众可得到的美国政府规范ICD-GPS-200中定义。D的卫星位置部分实际上是使用在围绕地球分配的五个监视站提取的GPS卫星的距离测量计算的一个预测量。周期地,通常是日常的操作,GPS控制部分把每个卫星的预测导航数据和估计校正量上载到星载原子钟。
卫星导航数据包括GPS日历,用于预测未来多周的每一GPS卫星的位置和速度。普通GPS接收机使用该日历数据、ICD-GPS-200中定义的算法和标准线性方程式求解技术来计算每一GPS卫星的位置和速度,并且预测接收机将发现这颗卫星的信号的卫星所在预期范围(PRN码相位)和多普勒频率。
因为所有的卫星广播是在相同的载波频率,每一个卫星航向信号必须能够以最小的干扰从其它信号中使用该频率。在一个称之为码分多址(CDMA)的方法中,这一点是通过认真选择PRN码具有尖锐(1个码片宽)自动相关峰值的PRN码实现的,以便实现码同步并且实现在整个频带上的均等扩展。对每一卫星来说该C/APRN码是唯一的并且取自称之为黄金码的一族码。该GPS C/A码被形成作为两个最大二进制码序列(G1和G2)的乘积(即模2加),每一序列是1023比特长。通过相对于G1移位该G2寄存的起始状态而产生这种黄金码族。根据两个判据,1023个可能的黄金码的32个码被选择用于GPS卫星:在该码中1和0的数目必须准确地相差1(即码被均衡),并且任何两个C/A码之间的互相关联不大于65/1023,或-23.9dB(归一化到1的自动相关峰值)。这种互相关联不敏感(immunity)被称作黄金结合,并且表示在具有完全相同频率的相等强度的C/A编码信号之间的最大干扰。这种PRN信号设计实现满意的GPS系统的CDMA操作,即多达32颗的卫星共享相同的广播频带,假定接收的GPS信号的功率不大于在通常情况下的该黄金结合。
黄金码结合可适用于具有相同载频的信号。然而,由于轨道中的卫星运动和接收机的移动引起的多普勒频移,该GPS卫星信号的接收频率通常从标称的1575.42MHz的L1载频偏移达±5KHz。相对于任何单一卫星,其它卫星的频率可能有大到±9KHz的差。
如果信号被多普勒频移,则该强/弱互相关联问题将更糟。如先前提到的那样,通过形成针对两个顺序之间的所有的1023个可能的时间偏移的最大二进制码序列(G1和G2)中所选择的一对的模2加而产生C/A码的黄金码族。二进制码的互相关联(两个信号的相乘)等效于码的模2加,因为±1值的乘积与二进制0、1的模2加具有一对一的对应关系。因此,黄金码族的两个多普勒频移项的互相关联减小到每一最大序列与自身的模2加,由另一模2加跟随。一个最大序列的偏移和相加的性质意味着一个最大序列与该同一个最大序列的偏移的模2相加将产生该同一个最大序列的另一个偏移。因此,黄金码族此两个多普勒频移项的互相关联产生同一个黄金码族的另一项。已经发现,这些产生的黄金码不是该C/A族的项,并且可以具有超过该C/A码设计限度的互相关联性。
没有具有很不同的载频的多普勒频移C/A码的互相关联干扰的闭合形成分析是已知的。相反,使用仿真的方法来分析有关C/A码互相关联的多普勒频移效应。该仿真或者产生两个期望的频偏码并且直接计算该互相关联性,或产生被调节用于频偏的每一码的傅里叶变换,并且计算该变换的互相关联性。已经发现,对于±9KHz的多普勒范围来说,GPS C/A的最糟情况的互相关联性是-20.9dB。当两个卫星信号之间的相对多普勒频移在1KHz的整数倍范围时,出现这种最糟情况。
当该频差是1KHz的一个整数倍时,虽然多普勒偏移增加强/弱信号的电平,但是当该多普勒频移不是1KHz的倍数时,频率衰减将降低该互相关联效应。在被用于信号检测或信号跟踪之前,该GPS接收机针对一定的时间长度积分(取和)该同相和正交(I、Q)的测量数据。如果该积分的信号包含一个频率误差,则该累加过程将通过公知的sin(x)/x函数降低该信号的表面强度,其中x是出现在积分周期上的弧度角表示的相位旋转的一半量(注意,随着x靠近0,sin(x)/x的极限是1)。因此,如果该复制的弱信号和干扰强信号之间的多普勒差值是500Hz,并且I、Q积分时间是1ms,则x等于π/2弧度角,sin(x)/x等于2/π,而干扰被衰减大致4dB。
结果是,如果一个卫星的信号强度高出第二颗卫星的信号强度接近20.9dB,则可能出现强/弱信号互相关联的问题。在此条件下,采集搜索可以检测该该强卫星的互相关联性的谱线,而不是检测弱卫星的自动相关性谱线。
该种GPS系统的设计假设其接收机在户外操作,具有直达所有的卫星的线路。在此情况中,C/A码提供足够的抵消强/弱信号互相关联的保护。然而,只要接收机在户内或在树冠遮盖下移动,某些信号将会变得严重衰减,而其它信号继续以正常信号强度接收。在这种情况中,黄金码互相关联峰值的操作上的严重性在于造成难于区别一个弱的GPS信号和相对较强的GPS信号的互相关联。一个错误的识别可能引起该GPS接收机计算的纬度、经度和高度中的大误差。
配备SPS的GPS接收机将在任何给定时间从多达十二颗的卫星接收L1航向信号,所有的L1航向信号在同一个载频上多路复用,每一个由其自身拥有的C/A PRN黄金码调制。接收机必须能够从这个复合载波信号识别并提取各个卫星的信号,随后处理这些信号的每一个,以便恢复包含在其中的信息。这些卫星的每一颗都具有与每一颗其它卫星信号干扰的可能。在最坏的情况中,当来自单一弱卫星的信号和多颗强卫星的信号被同时地接收时,该弱卫星信号可能会具有来自每一强卫星信号的严重互相关联干扰。
当一个GPS接收机被首先接通时,最好只大致了解其定位、本机振荡器偏移(将作为对全部卫星共同的一个多普勒频率偏移出现)和正确的时间。因此,该接收机必须执行一个贯穿全部可能C/A码相位的大部分和全部可能多普勒效应偏移的系统搜索,以便定位该卫星信号。在搜索来自任何相对强的卫星的强/弱互相关联的过程中,可以引起该接收机把来自该强卫星的一个互相关联谱线误认为是来自弱卫星的信号。
接收机已经起动以后,其能够使用天文年历数据和ICD-GPS-200算法来预测该C/A码相位以及所有卫星的多普勒偏移,在此时其只须针对该期望的卫星信号搜索C/A码相位和多普勒频率偏移的一个相对小的范围。但是,当在相对弱的卫星的搜索范围之内出现来自一个相对强的卫星的互相关联峰值时,这种强/弱互相关联问题依然存在。
一个普通GPS接收机包括一个天线,有选择地接收该载波信号而同时拒绝多路径信号和干扰信号;一个前置放大器包括带通滤波器,滤除可能的相邻频带中的高电平干扰信号,以及一个低噪声放大器(LNA),放大该载波信号;一个基准振荡器提供用于该接收机的时间和基准频率;一个由振荡器驱动的频率合成器;一个下变频器,把滤波的载波信号转换为中频(IF);一个IF部分,提供对带外噪音和干扰的进一步滤波、放大该信号到一个可使用的信号处理电平、并且有选择地把该IF信号下变频为一个基带信号;以及一个模-数转换器(ADC),把该信号取样和量化成同相(I)和正交(Q)成份。该ADC可以根据该接收机设计取样该IF或者该基带信号。
数字化的I、Q信号被随后馈送到1至12个或更多个跟踪信道。在那里与一个C/A PRN码副本相关,该C/A PRN码副本可以是按照要求利用移位寄存器内部地产生,或作为全套的预先计算C/A码的码片存储在存储器中。通过混频(相乘)该两个信号并且积分(取和)该产生信号的同相和正交分量的功率而实现该副本和接收信号的相关。通常,通过锁相环(PLL)、科斯塔斯(Costas)相位检测环路和/或延迟锁相环路(DLL)把此副本信号的载波和码的相位与接收信号校准。通过把该正交分量驱动至零而同时最大化该同相成份,该PLL和科斯塔斯环路保持该已接收和副本信号的相位一致。通过均衡以两个或更多个码偏移,例如提前和滞后即提前和准时测量的相关功率,该DLL保持C/A码的对准。每一个恢复的扩展频谱L1信号被随即馈送到接收机的信号处理部分,其中解调该信号,以便恢复信号载波和C/A及D码。这些信号又被提供到一个导引数据处理器,其中从D码计算每一颗被跟踪的卫星的位置,并且执行各种误差校正。误差源包括电离层和对流层延迟、多普勒效应、卫星和接收机时钟误差、设备延迟、噪音、以及由于信号被反射而因此多次接收但有轻微延迟产生的多路径误差。
由GPS接收机靠近地球表面接收的最大C/N0(在1Hz带宽中的信噪比)大致是55dB-Hz,允许附加的多路干扰。相对照,现有技术的GPS跟踪算法能够获得并跟踪具有低到24dB C/N0的GPS信号,并且未来技术进步有希望把此阈值进一步降低。因此,可用的GPS信号功率的范围是35dB或更高。假设最糟情况的强/弱互相关联C/A码的谱线是-20.9dB,则需要一种方法把该C/A码的识别提高至少10dB-Hz。
已有技术已经开发了一种通用的方法,在码定时、载波相位以及信号幅度是已知的条件下预测两个多普勒频移PRN编码序列的互相关联。该方案可以总结为按照计算集中的维特比算法的未知数据比特的最佳最大可能性解调。在实际意义上,这种最佳解调能被看作等效于以足够的引入延迟来实现强信号的消除,以便估计具有低差错率的该强信号的未知数据比特。该通常方案假设的是一个理想信道,但是CDMA近-远问题的一种实际方案还必须对付多路径传播的影响。
因此,根据上面的情况,本领域技术人员已经认识到需要在CDMA编码的扩展频谱信号中增加该强/弱信号的鉴别。本领域技术人员还认识到,开发适合于GPS的SPS的这样一种方法将是重要的。本发明就是满足这些以及其它要求。
本发明概要
所需要并且尚未得到的是一种用于消除一个强编码扩展信号对于弱编码扩展信号的影响的方法,该影响即为CDMA的所谓的近/远或强/弱问题,需要该方法能在现有的系统中执行而不超出系统容量限制。
本发明的方法实现了强信号对于弱信号影响的传递相关性的消除,并且能在几乎任何多信道接收机中实施,对整个信道的容量负荷仅有很小的添加。该实现的校正的弱信号把CDMA接收机的作业范围延伸到传统上的难区,例如建筑物围绕区域以及建筑物之内、或在一个森林地带的树冠下。
总的来说,该方法包括跟踪在一种多信道CDMA接收机,例如一个GPS接收机中的一个或多个强信号。使用关于可得到的信号源的信息,接收机可以分类被认为将要出现但当前没有被作为弱信号跟踪的任何信号源。可以通过消除所有的强信号对于弱信号的互相关联影响来跟踪这些弱信号。这是通过把该多信道接收机的一个信道设置到每一弱信号的预测频率和码相位实现的。来自这一信道的测量将包含任何强信号与期望弱信号的互相关联性。该互相关联性能够通过把强和弱信号信道的编码序列互相关联而计算。因为强信号是正被跟踪的信号,所以其幅度和相位是已知的。最终,如先前讨论的那样,当信号之间的相关多普勒效应是1000Hz的整倍数时,该互相关联性具有最大峰值。通过由强和弱信号之间的频率差引起的衰减定标每一跟踪的强信号并且乘以计算出的互相关联性,能够估计并且因此消除该强信号对于该弱信号的影响。为了以PLL、DLL和科斯塔斯环路实现弱信号的两个载波和码的跟踪,必须针对至少两个基准码偏移重复该处理,例如针对提前和滞后或提前和准时的两个基准码偏移重复该处理。
弱信号的信号检测能够以两个方式之一控制。最简单的方法是仅当强和弱信号之间的增量频率(实际接收频率中的差值)提供该强信号与该弱信号的互相关联的足够衰减时才执行信号检测。更完整、但较慢且更复杂的方法是在多普勒频率和可能的码偏移的适当范围上搜索,把消除该互相关联的强信号的方法用于全部可能的多普勒和编码偏移。
从下面结合附图的详细描述,本发明其它特征和优点将变得显见,附图以实例的方式示出了本发明的原理。
附图的简要描述
图1是描述本发明方法的主要步骤的流程图。
详细描述
由扩展频谱、例如使用在CDMA中的PRN码族提供的强/弱或近/远信号的隔离取决于该系列的各种码项之间的互相关联性。在GPS的情况下,在同一频率(或多个编码重复率,在此情况中是1KHz)的两个信号的隔离是大约21到23dB。如果两个信号的相对强度差超过此限度,则就不能仅使用该扩展码识别该弱信号。如果该弱信号是将要跟踪的信号,则必须应用一种消除这强信号影响的方法。
如上所指出,在C/AGPS信号的情况下,当相对强和弱信号之间的相关多普勒频率偏移是1KHz的整数倍时,该互相关联效果在其最大值。
解决跟踪一个弱信号问题的一个通常方案是在存在已经生成的一个较强扩展频谱信号的情况下扩展频谱信号。根据的前提是,该强信号干扰的所有方面都能被测量或计算,以便从该弱信号中除去该强信号。该方案能够在任何多信道接收机中实现,条件是该多信道接收机具有能力来控制信道的频率和相位以及选择期望的扩展码并且设置该码的相位位置。该接收机通常采用两个信道,一个信道跟踪该弱信号而一个信道跟踪该干扰强信号。然而,如果例如该强信号的功率、码相位和频率的特征可以被获得或通过另外的装置精确估算的话,则不需要用于跟踪该强信号的信道。
如图1所示,在步骤10首先获取该强信号,例如通过在接收机的第一信道中迟踪。随着载波信号和扩展码的相位,该信道提供该强信号的信号强度的测量。另外,信道可被用于跟踪另外的强信号(流程图中没有示出)。
根据数据信息D,通过已有技术中的公知方法,在步骤20,连同其接收频率和信号相位一起预测该弱信号的扩展码的码相位。接收机中的第二信道专用于接收该复合载波信号并且在步骤30跟踪该预测的弱信号成份。
该第二接收器信道以预测的频率和信号相位把输入信号与第二编码相关。产生的同相和正交(I,Q)测量既包括弱信号又包括强信号,每一个由其唯一的码扩展。通过倍增第二信号的复制码Code2R与输入信号相关,得出乘积Code2R×(weak2×Code2+StrongX×CodeX+...),其中weak2是弱信号2的功率,Code2是广播该弱信号2的卫星2的实际编码,StrongX(X=1,3,4,...)是强信号X的功率,而CodeX是包含在信号中的卫星X的实际编码。乘积Code2R×Code2是接收的码2和复制码2的自动相关。如果复制码与接收码对准,则该自动相关函数具有一个1的值。复制码2与码X的互相关联性(Code2R×CodeX)被随后以步骤40计算,以便从该复合信号中去除这种互相关联性。
Codel和Code2都是一个PRN码族的组成部分,并且其自动相关性和互相关联性质是已知的。因此,通过以Codel的每一比特简单地相乘Code2的对应(时间)比特而生产它们的互相关联值,有可能以其分别相位计算这两个码的互相关联性。由于有可能存在该两个码之间相关多普勒频率偏移,所以该码的相位将在时间上逐个变化,并且产生一个新的互相关函数。对于GPS系统来说,通常遇到的最大增量码多普勒效应是大约9KHz,等效于6个码片/每秒(1540载波周期/码片),因此互相关联值的最大重新计算速率是大致6次/每秒。
如前面指出,最大互相关联值出现在以1000Hz间隔出现的峰值的零频偏的位置。随着频偏离开零,出现该互相关联的衰减。此衰减遵循公知的sin(x)/x曲线。如果10ms测量被用于跟踪或采集,则该衰减因数将等于sin(Δfreq×π/100Hz)/(Δfreq×π/100Hz)。这将以大约75Hz的增量频率产生-10dB的衰减。该sin(x)/x曲线的其它局部峰值(即局部最小值衰减)将分别出现在具有-13.5和-18dB的衰减的150和250Hz之处。这将意味着,对于一个期望的10dB强信号的抑制来说,仅需要考虑该sin(x)/x函数的第一正弦的半周;但是,倘若期望进一步的抑制,可以考虑整个曲线。
随后的步骤50将针对每一强信号计算该强信号幅度和计算的频率以及时域(码相位)互相关联性的乘积。在步骤60通过从复合信号中减去此乘积而最后提取该弱信号。因此提取的弱信号随后在接收机电路中按照已知技术处理。
通过测量每一强信号自身的各个接收机信道或通过独立装置的估计而获得每一强信号的同相和正交幅度(I,Q)。因为该强信号是由接收机的锁相环动态跟踪,所以该强信号的相位被假设为接近零弧度角,并且因此几乎全部的信号功率是在该同相部分中。
以第一码Code1调制的包括强信号S1的一个信号与以第二码Code2调制的一个弱信号w2取和,产生(S1×code1+w2×Code2)。两个信号的取和与该第二码Code2R的一个副本相关,生产∑{Code2R×(S1×Code1+w2×Code2)},其中取和∑包括使用调制该弱信号w2的PRN码的所有的码片。一个码与自身的自动相关是1,所以前面的方程式能够重新写为∑{S1×code1×Code2+w2}能够知道,为了获得w2,必须除去S1×code1×Code2。由于已知Code1和Code2,所以能够容易地计算其互相关联性。这将实现通过独立地跟踪在一个独立信道上的强信号、或通过任何其它合宜的手段而完成S1的值的估计。如果强信号S1和弱信号w2在相同的频率,则将完全能够得到S1×Code1×Code2的计算值。然而,由于多普勒效应以及前面列举的其它因素,这两个信号是以不同频率接收的。已知该互相关联的强度随在频率之间的差值以sin(x)/x关系改变。因此必须根据该强和弱信号之间的频率差值计算一个衰减因数,并且将其用于计算出的互相关联性。此外,如果存在一个以上的强信号,则必须针对每一个强信号计算一个衰减因数。
码相关的互相关联因数的计算
从PRN代码发生器的已知相关状态计算该互相关联因数的码相关部分,以便预测单位功率的强信号和零频率偏移以及弱信号之间的互相关联性。这因数由对应于强信号的幅度相乘,并且在其从该复合信号减去之前针对频率衰减作调整。
用于调制该PRN信号的各种黄金码都是从两个编码序列G1和G2获得,其中在G2已经根据该所选的黄金码相对于G1已经偏移某些比特数目以后,通过XOR操作结合这两个序列的比特。如在本说明书其它部分指出的那样,使用二进制数字的一个XOR操作在数学上等效于相乘±1。这使得下面以±1的乘积表示该方程式,而实际上的实施可以是二进制数字的XOR操作。
两个C/A码之间的相关性通常能够被表示成:
∑Sat1G1(1)×Sat1G2(I)×Sat2G1(I-offset)×Sat2G2(I-offset)×e-jΔθl
其中
I=从0到1022的取和指数范围
Sat1G1(I)=卫星1的在状态I的G1编码器码片的值。可能的值是±1
Sat1G2(I)=卫星1的在状态I的G2编码器码片的值。可能的值是±1
Sat2G1(I)=卫星2的在状态I的G1编码器码片的值。可能的值是±1
Sat2G2(I)=卫星2的在状态I的G2编码器码片的值。可能的值是±1
offset(偏移)=单位码片中的卫星1和2之间的时间差
Δθ=以弧度为单位的在卫星1和2之间的每一码片的相位变化
应该指出,当差值I-偏移小于0时,把1023加到该差值,以便保持该值在0到1022的范围。换言之,返回编码器码片状态的该函数的范围局限于0到1022的范围。
用于计算该1023个逐个比特相关所需的计算时间能够通过利用标准的CPU指令而加速,以单一CPU指令执行8、16或32比特的XOR操作。下面将示出并行计算8个码片的方法。本领域技术人员将立即认识到,本方案可被容易地修改,以便适应每一CPU XOR操作的其它合适的比特数。
G1和G2的1023个状态被线性地存储在永久性存储器中。因此有可能通过计算期望码片的地址和为了对准该地址所要求的移动而利用单个CPU装入指令快速收集器8、16、32或其它合宜的比特数。32比特是一个尤其合宜的数目,因为31除尽1023。因此该最佳实施例同时读出32比特,并且针对复盖该C/A码的1023码片的33个间隔的每一个,同时使用31个32比特。该31比特的取和被分为4个部分:8、8、8和7比特,并且每一个7或8比特取和乘以e-jΔθt,其中针对每一个部分,I变化了7.75码片。取和的形式是:
∑(e-jΔθl×31×∑(Sat1G1(I×31+J)×Sat1G2(I×31+J)×
Sat2G1(I×31+J-offset)×Sat2G2(I×31+J-offset))+
e-jΔθ(1×31+7.75)×∑(Sat1G1(I×31+J+8)×Sat1G2(I×31+J+8)×
Sat2G1(I×31+J+8-offset)×Sat2G2(I×31+J+8-offset))+
e-jΔθ(1×31+15.5)×∑(Sat1G1(I×31+J+16)×Sat1G2(I×31+J+16)×
Sat2G1(I×31+J+16-offset)×Sat2G2(I×31+J+16-offset))+
e-jΔθ(1×31+23.25)×∑(Sat1G1(I×31+J+24)×Sat1G2(I×31+J+24)×
Sat2G1(I×31+J+24-offset)×Sat2G2(I×31+J+24-offset)))
其中
I=从0到32的外指数范围
J=针对开头三个取和的从0到7的内指数范围以及针对最后一个取和的从0到6的内指数范围。使用包含全部31个比特的一个32比特字以及使用逐比特XOR而并行计算内取和,以便执行该相乘和移位,并且相加以便取和该1比特乘积。
注意,在上述方程式中的G1和G2的全部相乘是通过比特XOR指令实施的。从准确计算的角度看,上面的算法最多错误-17dB,并且需要大约6000个CPU操作来完成。
计算的互相关联性的使用
按照要求,针对具有小频差的全部强和弱信号对计算码相关的互相关联系数,即针对全部能够引起强弱互相关联干扰的频差计算码相关的互相关联系数。在该最佳实施例中,强信号是具有C/N0>40dB的信号,而弱信号是具有C/N0<30dB的信号。由于码与相位跟踪环路使用10ms的I、Q测量的积分,所以最大的″重要″频差(模1000Hz)是90Hz。在该最佳实施例中,针对可以由跟踪和信号处理算法使用的每一个测量,计算针对每一可能的信号干扰对的码相关交互相关因数。例如,如果该跟踪环路使用的是提前、准时、和滞后的测量,则用于这些码校准的每一个的相关因数被计算并且存储在表格中。
这些表格仅需要以10Hz的速率更新,因为最大的多普勒差值是小于9KHz,即小于6码片/每秒。除了维持该互相关联表格之外,以10Hz速率计算由于频差引起的该互相关联的频率衰减。该衰减能够表示成:
频率衰减=sin(ΔFmod1000×π/100)/(ΔFmod1000×π/100)
其中
ΔF=以Hz为单位的强和弱信号之间的频差
Mod=对于-500Hz到+500Hz范围的模偏移
如果频差改变超过5Hz则需要重算该衰减。
强信号估计和消除
为了去除该互相关联性,需要估算该强信号的相位和幅度。本最佳实施例中使用的方法是跟踪在其自身拥有的专用信道上的强信号,并且收集在提取该弱信号的I、Q取样的完全相同间隔上的I、Q测量输出。用于跟踪该强信号的该复制信号的已知相位和频率是该强信号的实际相位和频率的一个最佳近似。此外,因为该强信号被锁相,所以该I测量的幅度提供该强信号幅度的一个良好近似。最终,该强信号数据比特D的双相位调制能够使得该强信号的相位每当该数据比特从I过渡到Q或从Q过渡到I时旋转180度。在该最佳实施例中,每当用于强信号的I测量的符号是负值时,则通过对该复制信号的相位相加180度而校正该强信号的相位。
每10ms可从指定用于跟踪该弱信号的信道得到一组新的I、Q相关数据。查验互相关联因数的表格,以便预测任何干扰强信号的出现。如果预测到强信号,执行下面的相减,以便除去该强信号的互相关联性。
第一码偏移=弱码状态-强码状态-强多普勒效应×ΔT-表格输入项0码状态
增量相位=弱载波相位-强载波相位-强多普勒效应×ΔT+增量KHz×强码状态
第一相位=第一相关相位+增量相位
第二相位=第二相关相位+增量相位
第一幅度=第一相关幅度+第一码偏移部分×强I×频率衰减
第二幅度=第二相关幅度×(1-第一码偏移部分)×强I×频率衰减
校正的弱IQ=弱IQ-第一幅度×e-j第一相位-第二幅度×e-j第二相位
其中
弱码状态=最后输出到弱信号信道的码状态
强码状态=最后输出到强信号信道的码状态
强多普勒效应=最后输出到强信号信道的多普勒效应
ΔT=在输出到弱和强信道之间的时间差值
表格输入0码状态=互相关联表格的第一成分的码状态差
弱载波相位=最后输出到弱信号信道的载波相位角
强载波相位=最后输出到强信号信道的载波相位角
增量KHz=弱和强信道多普勒效应之间的差值的1KHz的最接近的整数倍。以KHz为单位。
第一相关相位=针对由第一码偏移指示的码片,在互相关联表格中的相位输入
第二相关相位=针对由第一码偏移指示的码片+1码片,在互相关联表格中的相位输入
第一相关幅度=针对由第一码偏移指示的码片,在互相关联表格中的幅度输入
第二相关幅度=针对由第一码偏移指示的码片+1码片,在互相关联表格中的幅度输入
第一码偏移部分=在第一码偏移中的一个码片的局部
强I=来自强信道的I相关性的绝对值
频率衰减=由于频偏引起的衰减
弱IQ=来自弱信号信道的IQ相关性
校正的弱IQ=针对来自强信号的互相关联性校正的IQ相关性
通过适当地移动第一码偏移,例如每次移动半个码片,针对提前、准时和滞后相关器计算校正的弱IQ。这些修正的相关性被随即正常使用在针对弱信号的载波和码跟踪软件中。该算法至少把互相关联性衰减10dB而不衰减弱信号,并且针对每一可能干扰弱信号的强信号而重复该算法。
虽然已经示出和描述了本发明的一个具体形式,但是在不背离本发明精神和范围的条件下,显然能够作出各种修改。必须理解到,虽然已经依据对于GPS接收机的应用描述了最佳实施例,但是本发明的方法能被用于采用易受近-远问题的影响并且其中该干扰强信号能被以充分的精确度测量或估算的CDMA扩展频谱传输的任何其它通信系统。因此,本发明不打算受到除了所附的权利要求书之外的限制。
Claims (10)
1.操作扩展频谱无线信号接收机的方法,该接收机具有至少第一和第二信号跟踪信道,以便增加在同一载波频带上广播并且分别由第一和第二码调制的第一和第二接收信号之间的干扰识别,该码的每一个具有已知数目的码相位,本方法包括步骤:
(a)接收包括第一和第二接收信号的第一复合信号;
(b)操作该第一信道跟踪该第一接收信号并且测量其幅度、码相位和接收频率;
(c)选择该第二接收信号的接收频率和码相位;
(d)以该第二接收信号的所选码相位和接收频率操作第二信道,以便接收包括该第一和第二信号的第二复合信号;
(e)计算该第一接收信号与该第二接收信号的预测码和频域互相关联性;
(f)把第一接收信号的幅度与预测的互相关联性相乘,以便计算在该第一和第二接收信号之间的干扰;和
(g)从该第二复合信号减去该干扰,以便提取包括其接收频率、幅度和码相位至少之一的第二接收信号。
2.根据权利要求1的方法,包括进一步的步骤:
在步骤(c)中,执行进一步的步骤:
针对该第二接收信号选择接收频率和码相位的一个范围;
在分别的所选范围之内选择一个初始第二码相位和接收频率;和
选择一个码相位增量值和一个接收频率增量值;
针对该第二接收信号的最初第二码相位和初始接收频率,执行步骤(d)-(f);
执行步骤(g),以便提取该第二接收信号的幅度;
针对在各个范围中的所有的第二码相位和接收频率值重复步骤(d)-(g),该范围从分别的初始值分别偏移了码相位递增值和接收频率递增值的一个整数倍;和
选择该第二接收信号码相位和接收频率,在该相位和频率,该提取的第二接收信号具有用于在第二信道中进行跟踪的最大幅度。
3.根据权利要求2的方法,包括进一步的步骤:
在步骤(c)中,执行进一步的步骤:
选择频域互相关联性的门限值;
在步骤(e)中,把该频域互相关联性与该门限值比较;和
如果该互相关联性大于这门限值,则直接针对第二接收信号的接收频率的一个不同值执行步骤(d)。
4.权利要求3的方法,其中该互相关联性的门限值是大约10dB。
5.操作扩展频谱无线信号接收机的方法,该接收机具有至少第一和第二信号跟踪信道,以便增加在同一载波频带上广播并且分别由第一和第二码调制的第一和第二接收信号之间的干扰识别,该码的每一个具有已知数目的码相位,本方法包括步骤:
(a)接收包括第一和第二接收信号的第一复合信号;
(b)操作该第一信道,以便跟踪该第一接收信号并且测量其幅度;
(c)预测该第一接收信号的第一码相位和接收频率;
(d)选择该第二接收信号的第二码相位和接收频率;
(e)以该第二接收信号的所选码相位和接收频率操作第二信道,以便接收包括该第一和第二接收信号的第二复合信号;
(f)计算该第一接收信号与该第二接收信号的预测码和频域互相关联性;
(g)把第一接收信号的幅度与预测的互相关联性相乘,以便计算在该第一和第二接收信号之间的干扰;和
(h)从该第二复合信号减去该干扰,以便提取包括其接收频率、幅度和码相位至少之一的第二接收信号。
6.根据权利要求5的方法,包括进一步的步骤:
在步骤(d)中,执行进一步的步骤:
针对该第二接收信号选择接收频率和码相位的一个范围;
在各个所选范围之内选择一个初始第二码相位和接收频率;和
选择一个码相位增量值和一个接收频率增量值;
针对该第二接收信号的起始码相位和初始接收频率,执行步骤(e)-(g);
执行步骤(h),以便提取该第二接收信号的幅度;
针对在各个范围中的所有的第二码相位和接收频率值重复步骤(e)-(h),该范围从分别的初始值分别偏移了码相位递增值和接收频率递增值的一个整数倍;和
选择该第二接收信号码相位和接收频率,在该相位和频率,该提取的第二接收信号具有用于在第二信道中进行跟踪的最大幅度。
7.根据权利要求6的方法,包括进一步的步骤:
在步骤(d)中,执行进一步的步骤:
选择频域互相关联性的门限值;
在步骤(f)中,把该频域互相关联性与该门限值比较;和
如果该互相关联性大于这门限值,则直接针对第二接收信号的接收频率的一个不同值执行步骤(e)。
8.权利要求7的方法,其中该互相关联性的门限值是大约10dB。
9.操作扩展频谱无线信号接收机的方法,该接收机具有至少一个信号跟踪信道,以便增加在同一载波频带上广播并且分别由第一和第二码调制的第一和第二接收信号之间的干扰识别,该码的每一个具有已知数目的码相位,本方法包括步骤:
(a)获取该第一接收信号的幅度、码相位和接收频率;
(b)选择该第二接收信号的接收频率和码相位;
(c)以该第二接收信号的所选码相位和接收频率操作该跟踪信道,以便接收包括该第一和第二接收信号的第二复合信号;
(d)计算该第一信号与该第二接收信号的预测码和频域的互相关联性;
(e)把第一信号信号的幅度与预测的互相关联性相乘,以便计算在该第一信号和第二接收信号之间的干扰;和
(f)从该第二复合信号减去该干扰,以便提取包括其码相位、接收频率和幅度的第二接收信号。
10.权利要求9的方法,其中该接收机包括一个附加信号跟踪信道,并且在步骤(a)中包括进一步的步骤:
在该附加信道中接收包括该第一接收信号的一个附加复合信号;和
操作该附加信道,以便跟踪该第一接收信号并且测量其幅度、码相位和接收频率。
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- 2000-11-14 EP EP00992803A patent/EP1238485B1/en not_active Expired - Lifetime
- 2000-11-14 DE DE60022901T patent/DE60022901T2/de not_active Expired - Lifetime
- 2000-11-14 KR KR1020027007651A patent/KR100676780B1/ko not_active IP Right Cessation
- 2000-11-14 WO PCT/US2000/042171 patent/WO2001047171A2/en active IP Right Grant
- 2000-11-14 JP JP2001547785A patent/JP4738692B2/ja not_active Expired - Fee Related
- 2000-11-14 AU AU47078/01A patent/AU4707801A/en not_active Abandoned
- 2000-11-14 CN CN00817124A patent/CN1409904A/zh active Pending
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2001
- 2001-06-20 US US09/886,671 patent/US6707843B2/en not_active Expired - Lifetime
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Cited By (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
CN103176189A (zh) * | 2013-03-08 | 2013-06-26 | 浙江大学 | 高灵敏度卫星导航接收机的远近效应抑制器及其方法 |
CN109474288A (zh) * | 2019-01-14 | 2019-03-15 | 上海创远仪器技术股份有限公司 | 基于反相抵消机制提高接收机动态范围的电路结构 |
CN109474288B (zh) * | 2019-01-14 | 2024-03-15 | 上海创远仪器技术股份有限公司 | 基于反相抵消机制提高接收机动态范围的电路结构 |
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US7116704B2 (en) | 2006-10-03 |
EP1238485A4 (en) | 2003-05-07 |
EP1238485B1 (en) | 2005-09-28 |
KR20020068055A (ko) | 2002-08-24 |
ATE305677T1 (de) | 2005-10-15 |
KR100676780B1 (ko) | 2007-02-01 |
EP1238485A2 (en) | 2002-09-11 |
JP2003518819A (ja) | 2003-06-10 |
WO2001047171A2 (en) | 2001-06-28 |
AU4707801A (en) | 2001-07-03 |
US6707843B2 (en) | 2004-03-16 |
WO2001047171A3 (en) | 2001-11-15 |
JP4738692B2 (ja) | 2011-08-03 |
US20050032513A1 (en) | 2005-02-10 |
DE60022901D1 (de) | 2006-02-09 |
US20010046256A1 (en) | 2001-11-29 |
US6282231B1 (en) | 2001-08-28 |
DE60022901T2 (de) | 2006-06-29 |
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